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1/14 AN1621 application note 1 introduction a typical off-line isolated switch-mode power supply has the controller located on the primary side of the trans- former, whereas the output voltages to be controlled and the housekeeping functions are located on the sec- ondary side. usually the voltage feedback signal is transferred to the primary controller by using an optocoupler or a transformer. we want to propose here an asymmetrical half bridge fo rward converter with the controller located on the sec- ondary side. this solution offers some advantages: ? direct use of the on-board voltage reference ? no need of the opto feedback, with its temperature and ageing gain dependence ? available on-board housekeeping functions eliminat es the need of an additional dedicated device ? negligible extra cost on the gate drive transformer to satisfy safety requirements the controller operating on the secondary side requires a specific concept for the start-up sequence, realised here with a very simple, low consumption and cost effective solution. 2 power supply description 2.1 topology considering the 300w of output power delivered to the load, the most appropriate topology is the asymmetrical half bridge converter. this topology requires two power mos transistors with a voltage breakdown equal or little higher than the max. rectified mains voltage (thanks to the two clamping diodes), with a proper r dson to reach the target efficiency. fig. 1 shows the complete schematic diagram of the 300w power supply. 2.2 start-up circuit as already mentioned in the introduction section, the controller is located on the secondary side of the power transformer, and for this reason, a start-up circuit has to be provided for a correct system activation. the start-up circuit is based on a diac sending a train of controlled pulses to the low side drive section; the float- ing drive section is energised by the second secondary gate drive transformer winding . as soon as the l5991a wakes-up, it generates a pwm signal enabling the start-up circuit. 2.3 gate driver the two power mosfets, t1 and t2, are driven by a small transformer designed to satisfy the isolation safety requirements and fast switching times. for optimum magnetic coupling (required by the high switching frequency operation) and minimum number of turns, a high permeability core has been selected. 300w secondary controlled two-switch forward converter with l5991a AN1621/1104 rev. 2
AN1621 application note 2/14 figure 1. schematic diag ram of 300w power supply 10 r15 10 13 r8 430 12 11 r11 1.2k 24 c11 4.7 f r5 4.7k c12 1nf l5991a v ref rct 7 ss c16 390pf isen out sgnd pgnd 10t c5 1 f p1 10k c14 470nf tsf2 10t 15 16 dcl c10 2.2nf r1 3.6 d9 18v c29 3 3nf c7 100 f l6 9t d5 1n4148 98 vc vcc 5 6 fb comp c9 1nf r7 80k r12 6.2k 14 dis st-by r10 270 c18 100nf r9 3.3k c2 1000 f c3 1000 f c4 1000 f l1 24t 77930 24v 13a vo r30 180 d18 1n4148 t5 bc327 r24 30k t1 stw14nk50 c17 100nf d19 1n4148 10t r23 180 d16 1n4148 t4 bc327 r19 30k c15 100nf d17 1n4148 10t c26-27 220 f 400v emi 220vac bridge kbubj d12 1n4148 c25 100nf t3 bc337 c28 330pf r20 7.5k diac db3 1t d1 d15 stth1l06 d14 stth1l06 d13 stth1l06 r14 22k c3 100pf 32t tsf1 d3 in4148 d2 in4148 50t c6 1nf r2 10 r16 360k 50t 1t t2 stw14nk50 c30 2.2nf y1 byv52-200 in455_mod
3/14 AN1621 application note table 1. power supply specification the core is e20/10/5-3c85, 10 turns/winding, no air-gap. a high permeability core minimises the magnetising current to maintain a correct operation far away of the core saturation point, at maximum duty cycle and high core temperature. the proposed gate drive circuit allows the use of the 1:1:1 turns ratio drive transformer; moreover, a controlled drain current rise time (by r15) and a fast fall time are achieved. a very short min. ton pulse give the possibility to stabilise the output voltage at max. mains and min. load. 2.4 power mos selection the two-switch topology allows to use the power elements with a voltage breakdown equal to the max. rectified mains voltage. the mosfet used here is the stw14nk50; this device, with 500v of bvdss give us also some safety margin. the basic parameters of the stw14nk50 are listed below: rdson (25c) = 0.38 ? max., at id = 7a (0.76 ? max. at 100c) coss = 300pf typ., qg = 75nc typ., package in to-247 at min. supply voltage and max. load current, the conduction losses for each transistor are: pcon = i 2 rmsp rdson(100c) = 2.23 2 0.76 = 3.8w where the effective irmsp 2 is calculated in the power trans- former section. estimating in about 3.2w switching and parasitic losses, the total power losses of each transistor are about 7w. considering 100c of maximum operating junction temper ature (at 40c of ambient temperature) and a thermal resistance junction-heatsink of 0.66c/w, a heatsink of 4c/w is required to dissipate both the transistors. 2.5 current sense considering the output current rating of 13a continuous , it's our opinion that a current transformer for current sensing is the best approach for maximising the efficiency, reliability and internal ambient temperature in case the power supply has to be housed in a plastic box. due to the constant current limiting requirements, as shown in fig2, a couple of current transformers have been used; one transformer is sensing the current flowing in to d1 (in conduction when t1 and t2 are on) and the second one is sensing the current flowing into d5, recirculation diode. oring the two transformers by d2 and d3, and closing the loop with a proper impedance value, r1, we realise a voltage signal reproducing exactly the inductor current shape. the current sensing loop is closed by r1 selected according the transformers turns. two small toroid ferrite cores (41005-tc, magnetics, f material, 3000 ) have been used, with 50 turns. r1 is defined by: where: 1v is the nominal threshold voltage of the current sense. ipk is the inductor peak current( considering a 20% of current ripple, ipk = io + ? i/2 = 13 + 1.3 = 14.3a) symbol description parameter v i input voltage 220vac (176 vac to 265 vac) // 50hz v o output voltage 24v % (ripple voltage <1%) i o output current 13amax. continuous, 0.5amin i l limiting current constant type till ground p o output continuos power 312w f s switching frequency 200khz target efficiency (full load) = 90% (from mains to output) r1 50 1v ? lpk ------------------ =
AN1621 application note 4/14 figure 2. output current limiting charac teristic using two current transformer the calculated value is r1=3.5 ? fig. 2 shows the constant current charac teristic using two current transformers. the difference from the two current values, at output short-circuit and at current limiting intervention, is propor- tional to the half of chocke ripple current.a choke with higher value or higher switching frequency, can reduce this difference. if constant current feature is not requested to be constant till the output is reaching zero v, one single current transformer can be used. the new limiting current characteristic and the schematic diagram are shown in fig 3a and 3b:in order to reduce the peak current before hiccup intervention, an offset can be superimposed to the current sensing circuit to an- ticipate the hiccup limiting current intervention, by using the additional network shown below ( figs. 4a and 4b): using this solution, also the ? i/ ? vo is higher, reducing the difference from the intervention point and the short circuit current values.this solution can be used with two current transformers too. figure 3. output current limiting characteristic using one current transformer 8101214io 0 5 10 15 20 25 vo d96in456 a: output current limiting characteristic using one current transformer. b: limiting current schematic diagram 468101214io 0 5 10 15 20 25 vo d96in457 l1 d1 c6 10t r2 d2 r1 3.5 ? r8 430 ? c16 isen d5 13 l5991a d96in458_mod 1t 50t 330pf
5/14 AN1621 application note figure 4. output current limiting characteristic with foldback 2.6 output diode selection the reverse voltage of the output diodes is given by the formula: vr = vin max/n = 375/3.3= 114v, where n is the transformer turns ratio. for a correct functionality a ultra-fast recovery diode is requested, mainly to limit switching losses and emi prob- lems. to calculate the conduction losses the following equation has been applied to the selected type, byv52-200. for the single diode the formula is: p = 0.7 i (av) + 0.0075 i 2 (rms) that, rearranged to take into account both the diodes, becomes: p = 0.7 i omax + 0.0075 i o2max = 10.4w where i omax = 13a. considering the to247 package (1.2 c/w total thermal resistance junction-heatsink), to ensure that the junc- tion temperature does not exceed 100 c at 40 c max. ambient temperature, the heatsink has to be dimen- sioned for about 4c/w. 2.7 power transformer design the forward transformer delivers energy from the primary to the secondary without any storage. the only consideration is with regards of the magnetising current, that has to be limited at a safety value far from core saturation. our core selection is based on ap, area product, defined as ap = aw ae. a: output current lim iting characteristic b: schematic diagram of the modified 468101214io 0 5 10 15 20 25 vo d96in459 13 r8 3k d96in460_mod l5991a v ref c16 isen 10t p1 r1 4 3k c2 c3 c4 l1 1 t d1 d5 d2 c6 r2 r11 1.2k 50t 5 fb 120k 2k hiccup limiting current thereshold.
AN1621 application note 6/14 the minimum ap to avoid saturation is: the ap necessary to limit the core temperature rise not above 30 c is where: p o = output power ? b = flux swing f = switching frequency = efficiency ke =4 10 -10 kh = 4 10 -5 . in this application (p o = 312w, = 90%, f = 200khz) and considering a maximum ? b = 200mt for saturation, the most stringent condition is determined by the formula related to the core temperature raise. the minimum ap required is 1.97 cm 4 . the selected core is etd39 (ap = 2.2 cm 4 ) in 3f3 material. assuming total losses of 1% of p o (3w), 2/3 (2w) for the core and 1/3 (1w) for the copper, the specific core losses are: in presence of such a power losses, the flux swing can be determined, for 3f3 material, by using the diagram of material, at 200khz : figure 5. specific power loss as a func tion of frequency peak flux density with frequency as a parameter or by using the empirical formula with a specific core losses of 173 mw/cm 3 the flux swing must not exceed 130mt the minimum primary turns is given by: ap 67.2 p o ? bf ? ? ? ----------------------- - ?? ?? 1.31 cm 4 = ap 235 p o ? f ? ---------------------- ?? ?? 1.58 kh f ke f 2 ? + ? () 0.66 cm 4 = p pfe ve --------- 2w 11.5cm 3 ----------------------- 0.173w/cm 3 == = 1 2 5 10 20 50 100 200 ? b(mt) 10 20 50 100 200 500 1k 2k 5k p tot (kw/m 3 ) d96in461 t amb =100 c 1mhz 700khz 400khz 200khz 100khz 25khz 3f3 b ? p kh f ke f 2 ? + ? ------------------------------------ - ?? ?? 0.416 0.173 410 5 ? 200 10 3 410 10 ? 200 10 3 ? () 2 ?? + ??? ----------------------------------------------------------------------------------------------------------------- - ?? ?? ?? 0.416 130mt == =
7/14 AN1621 application note where ton is in sec, ? b in tesla and ae in mm2. the turns ratio is defined as: where: dmax = 0.48 (maximum duty cycle): vf =1v (voltage drop of the output diode); vp = 0.5v (voltage drop of the output inductor). the primary and secondary turns number are np = 32, ns = 10. as for the wire selection, the skin effect is not negligible at this frequency. to reduce this effect, the wire diameter should not exceed two times the penetration depth: and therefore a wire diameter of 0.36mm (awg27) is suitable. however, to have a sufficient copper area, a cer- tain number of these wires will be twisted together. to define the numbers of wires this equation can be used: where: r = resistance of the wire per cm lw = winding length (cm) as to the primary side: the rms secondary current is 3.4 times the primary one. considering the copper losses equally dissipated be- tween primary and secondary windings, this yields 3 wires in parallel (npwire) for the primary winding and 10 wires (nswire) for secondary winding. the etd39 core has a winding area of 1.7 cm 2 but the real available space is approximately 40% (0.68 cm2): isolation requirements and fill factor must be taken into account. the total copper area is: atot = ais (npwire np + nswire ns) = 0.255 cm 2 that fits the window. the windings are interleaved: one layer of 16 turns (first half of the primary), one layer of 10 turns (the secondary) and another layer of 16 turns (second half of the primary). with this arrangement the transformer used in the app lication has a leakage inductance of 15uh (about 0.5% of the primary inductance, which is 2.7 mh). the magnetising current is: n p v inmindc t on max () ? b ? a e ? --------------------------------------------------- 10 4 ? 200 2.4 ? 0.130 125 ? ----------------------------- 29.5 == n0.9 v in min () d max ? v o v f v p ++ --------------------------------------- - ? 0.9 200 0.48 ? 2410.5 ++ ------------------------------ - ? 3.38 === 7.5 f ------- - 0.17mm == n wire rn turns i w i rms 2 p loss ------------- - ??? = i rmsp p o v inmindc d max ?? ------------------------------------------------------ 312 0.9 200 0.48 ?? ------------------------------------------ - 2.23a === i m v in min () t on ? l p --------------------------------- 200 2.4 10 6 ? ?? 2.7 10 3 ? ? --------------------------------------- 180ma == =
AN1621 application note 8/14 the primary peak current is: a good condition is when im 10% of ipk; in this case im is imposed lower than 6% of ip. 2.8 output inductor & auxiliary supply to calculate the inductor value, the max. current ripple has been set at 2.6a, 20% of iomax. the max. current ripple occurs at max. input voltage; in this operating condition the min duty cycle is: the inductor value is defined taking into account the current downslope which corresponds to the off-time of the power switches: toff max. = t (1-dmin) = 5 (1-0.22) = 3.9 s that yields: to realise the inductor, a magnetic's kool mm core 77930 has been used. since the inductor has to assure the requested value at maximum load current, the turns number must be calculated taking into account the roll-off of the initial permeability. this results in 24 turns, that leads to 90 h at zero load. we can take advantage of this non-linear characteristic to keep a good output regulation and stability when working at min. load current( around 0.5a for this case).from the output inductor it is drown also the energy necessary to supply the l5991a controller by introducing a 9 turns auxiliary winding. 2.9 input capacitor the input filtering capacitor has to be dimensioned in order to deliver the requested max. output power, at min. mains value, with a reasonable ripple voltage at 100hz. the design process has to consider also the rms current flowing into the capacitor ( a major reason of stress) and the voltage rating. the minimum capacitor value is defined by: where: vcp is the peak value at minimum mains voltage, 248v vcm is the minimum value at minimum mains voltage, 200v the chosen capacitors are 2 x 220 f - 400v eys 06. at this point the new vcm value is 215v. the conduction time is: i pk p o v inmindc d max ?? ------------------------------------------------- - 312 0.9 200 0.48 ?? -------------------------------------- - 3.22a === d min nv o v loss + () ? v inmax --------------------------------------- - 3.2 24 1.5 + () ? 375 -------------------------------------- - 0.22 === l o v o v loss + () t offmax ? i max ? -------------------------------------------------------- - 24 1.5 + () 3.9 10 6 ? ?? 2.6 ------------------------------------------------------- 39 h === c p o f 50 v cp 2 v cm 2 ? () ?? ------------------------------------------------------- 312 0.9 50 248 2 200 2 ? () ?? ------------------------------------------------------------ 310 f == = t c cos 1 ? v cm v cp ---------- - ?? ?? ?? 2 f 50 ?? ------------------------------- - cos 1 ? 215 248 --------- - ?? ?? 23.1450 ?? ------------------------------ - 1.68 10 3 ? sec ? ===
9/14 AN1621 application note the input capacitor peak current is: the rms current that the capacitor has to substain is: considering the sensitivity to the temperature of the electrolitic capacitors, and the difficulties in fixing exactly the effective rms current value, for this particular part number, with an ambient temperature of 40c, the permit- ted temperature case is 85 c. the selected capacitors show a 1.4arms each of current capability at t case = 85c; that's enough to sustain our operating conditions. 2.10 output capacitors the output capacitor value is selected on the basis of the output voltage ripple requirement. this ripple is basi- cally due to the esr, since the capacitive component is by far lower. then it must be ensured that the total esr is below a maximum value of: where the ripple voltage has been fixed at 1% max. of vo. three capacitors eke 1000 f/35v (roe) have been paralleled, for a total esr of about 23mohm. 2.11 compensation network the power supply system can be simplified in two parts: power and feedback loop blocks. figure 6. closed loop block diagram the transfer function of the power block is: where: ? vo is the small signal output voltage ? vc is the small signal error amplifier output voltage i ch cv cp v cm ? () ? t c ---------------------------------------- 440 10 6 ? 248 215 ? () ?? 1.6 10 3 ? ? ------------------------------------------------------------- - 8.8a == = i rms i ch 2t c f 50 2t c f 50 ?? () 2 ? ?? ? 8.8 3.2 10 3 ? 50 3.2 10 3 ? 50 ?? () ? ?? 2 ? 3.32a == = i rmstot i rms50 2 i rms200khz 2 1.6 ---------------------------- + 3.32 2 2.23 2 + 3.76a === esr max v ripple ? i max ? --------------------- 0.24v 2.6a --------------- - 0.092 ? === g e/a g pw p ? v c ? v' o ? v o + + - int v ref (2.5v) d96in462 g pw s () v o ? v c ? ---------- n 3r s ? -------------- r o 1 s s z ------ + 1 s s p ------ + ---------------- ?? ==
AN1621 application note 10/14 sp is determined by the load (ro) and the output capacitance sz is determined by the esr and the value of the output capacitance n is the current transformer turn number rs is the sense resistance a load regulation of 1.5% has been imposed that corresponding to a maximum output voltage variation of 360mv. the power block dc gain is: considering a load variation between io=0.1a and io=13a, that corresponds to r o = 240ohm and r o = 1.84 ? , the ? v c excursion is: the e/a dc gain and the voltage divisor must be selected to fit this condition the output voltage divider usually sinks a current of about 2ma and the resistance values are: r11 = 12k ? p1+ r12 = 1k ? the r7 value is: it is necessary to introduce a pole at about one third of the zero frequency and c9 is: 2.12 overvoltage protection the voltage divider, r9 and r10, is providing for fixing the threshold for overvoltage protection intervention. with the selected values, the threshold is fixed at 30v. c18 is fixing the min. overvoltage time intervention to avoid ovp triggering from spikes. for an ovp function in tracking with the output voltage, in particular when v o is adjustable, it's enough to sub- stitute the r11 resistor with a voltage divider, and to connect the common point to pin14. 2.13 evaluation results fig 7 show the efficiency values obtained in different working conditions of input voltages and output currents. the efficiency is from ac mains to output dc voltage, at a measured switching frequency of 200khz. g pw v o ? v c ? ---------- nr o ? 3r s ? -------------- - == v c ? v o 3r s ?? n ------------------------- - 1 r omin --------------- - 1 r omax ----------------- ? ?? ?? ? 2.7v == v o ? v c ? ---------- 7.5 = r7 v o ? v c ? ---------- p1 r12 + () ? 80k ? == c9 1 2 f z ?? 3r7 ? --------------------- --------------------- - 1.8nf ==
11/14 AN1621 application note figure 7. efficiency versus output load current at min.,typ . and max. ac voltage. the efficiency is quite constant and equal or above 85% from 2a to full load. at min. load of 0.5a, it's 70% at 220vac. the total losses, for an amount of 3.6w are mainly due to coss of the power mos devices and transformer core losses due to the magnetizing current. fig 8. shows the load transient response, at nominal ac input voltage value and output load variation from 1a to 12a, in a couple of msec, and fig 9 shows the enlarged section at the moment of current load rise. fig 10. to be noticed the absence of output overshoot on the output voltage. figure 8. output voltage response to load transient 0 2 4 6 8 10 12 14 65 70 75 80 85 90 (%) d96in463 176vac 220vac 265vac d96in464 ch1: vo (100mv/div) ch2: io (5a/div) t: 1ms/div
AN1621 application note 12/14 figure 9. enlarged section of load tr ansient response at load current rise figure 10. turn-on at mains switch-on d96in465 ch1: vo (100mv/div) ch2: io (2a/div) t:100 s/div in466_mod ch1: vo (10v/div) ch2: l5991a self supply voltage c7=47 f (5v/div) t:100ms/div
13/14 AN1621 application note table 2. revision history date revision description of changes november 2002 1 first issue in edocd november 2004 2 new style-sheet in compliance to the "corporate technical pubblications desin guide" and labeling of schematic diagram
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