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rev.a, 10/08, wk page 1 of 14 MT-019 tutorial dac interface fundamentals by walt kester introduction this tutorial outlines some important issues regarding dac interface circuitry including the voltage reference, analog output, data input, an d clock driver. because adcs require references and clocks also, most of the concepts presented in this tutorial regarding these subjects apply equally to adcs. dac reference voltage there is a tendency to regard dacs simply as devices with digital i nputs and an analog output. but the analog output depends on th e presence of that analog input known as the reference, and the accuracy of the reference is almost always the limiting factor on the absolute accuracy of a dac. design tools such as the voltage reference wizard are useful is matc hing references to data converters. these tools a nd others are available on the design center portion of the analog devices' website. some adcs and dacs have internal references, while others do not. some adcs use the power supply as a reference. unfortunate ly, there is little standardi zation with respect to adc/dac voltage references. in some cases, the dc accuracy of a converter with an internal reference can often be improved by overriding or replacing the internal reference with a more accurate and stable external one. in other cases, the use of an external low-noise reference will also increase the noise-free code resolution of a high-resolution adc. various adcs and dacs provide the ca pability to use external refe rences in lieu of internal ones in various ways. figure 1 shows some of the popular configurations ( but certainly not all). figure 1a shows a converter which requires an exte rnal reference. it is generally recommended that a suitable decoupling capacitor be a dded close to the adc/dac ref in pin. the appropriate value is usually specified in the voltage reference data sheet. it is also important that the reference be stable with the required capacitive load (more on this to come). figure 1b shows a converter that has an internal reference, wher e the reference is also brought out to a pin on the device. this allows it to be used other places in the circuit, provided the loading does not exceed the rated va lue. again, it is important to place the capacitor close to the converter pin. if the internal reference is pinned out for external use, its accuracy, stability, and temperature coefficient is usually spec ified on the adc or dac data sheet.
MT-019 int ref int ref ext ref ref out ref out ref in int ref ext ref ref out/in r c c c adc/dac adc/dac adc/dac (a) (b) (c) int ref ref out ref in c adc/dac int ref ref out/in r adc/dac c (d) (e) ref in c adc/dac ext ref (f) figure 1: some popular adc/dac reference options if the reference output is to be used other place s in the circuit, the data sheet specifications regarding fanout and loading must be strictly obse rved. in addition, care must be taken in routing the reference output to minimize noise pickup. in many cases, a suitable op amp buffer should be used directly at the ref out pin before fanni ng out to various other parts of the circuit. figure 1c shows a converter which can use either th e internal reference or an external one, but an extra package pin is required. if the internal reference is used, as in figure 1c, ref out is simply externally connected to ref in, and dec oupled if required. if an external reference is used as shown n figure 1d, ref out is left fl oating, and the external reference decoupled and applied to the ref in pin. this arrangement is quite flexible for driving similar adcs or dacs with the same reference in order to obtain good tracking between the devices. figure 1e shows an arrangement whereby an external refere nce can override the internal reference using a single package pin. the valu e of the resistor, r, is typically a few k , thereby allowing the low impedance external reference to override the internal one when connected to the ref out/in pin. figure 1f shows how the exte rnal reference is conne cted to override the internal reference. the arrangements shown in figure 1 are by no mean s the only possible configurations for adc and dac references, and the individual data sheets should be cons ulted in all cases for details regarding options, fanout, decoupling, etc. page 2 of 14
MT-019 although the reference element itself can be either a bandgap, buried zener, or xfet?, practically all references have some type of output buffer op amp. the op amp isolates the reference element from the output and also provides drive capabil ity. however, this op amp must obey the general laws relating to op amp stability, and that is what makes the topic of reference decoupling relevant to the discussion. note that a reference input to an adc or dac is similar to the analog i nput of an adc, in that the internal conversion process can inject transient currents at th at pin. this requires adequate decoupling to stabilize the reference voltage. a dding such decoupling might introduce instability in some reference types, depending on the out put op amp design. of course, a reference data sheet may not show any details of the output op am p, which leaves the designer in somewhat of a dilemma concerning whether or not it will be st able and free from transient errors. in many cases, the adc or dac data sheet will recommend appropriate external references and the recommended decoupling network. a well-designed voltage reference is stable with heavy capaci tive decoupling. unfortunately, some are not, and larger capac itors actually increases the am ount of transient ringing. such references are practically usele ss in data converter applications , because some amount of local decoupling is almost always required at the converter. a suitable op amp buffer might be added between the reference and the da ta converter. however, there are many good references available which are st able with an output cap acitor. this type of reference should be chosen for a data converter application, rather than incurring the further complication and expense of an op amp. dac analog output considerations the analog output of a dac may be a voltage or a current. in either case it may be important to know the output impedance. if the voltage output is buffered, the output impedance will be low. both current outputs and unbuffered voltage output s will be high(er) impedance and may well have a reactive component specif ied as well as a purely resistiv e one. some dac architectures have output structures where th e output impedance is a functi on of the digital code on the dac?this should be clearly noted on the data sheet. in theory, current outputs should be connected to zero ohms at ground potential. in real life they will work with non-zero impedances and voltages. ju st how much deviation they will tolerate is defined under the heading "compliance" and th is specification shoul d be heeded when terminating current-output dacs. most high-speed dacs suitable for video, rf, or if, have current outputs which are designed to drive source and load-terminated cables directly. for instance, a 20-ma current output dac can develop 0.5 v across a 25- ? load (the equivalent dc resistance of a 50- ? source and load terminated cable). in most cases, single-supply high-speed cmos dacs have a positive output compliance of at least +1 v and a negative output compliance of a few hundred millivolts. page 3 of 14
MT-019 in many cases, such as the txdac? family, bo th true and complementary current outputs are available. the differential output s can drive the primary winding of a transformer directly, and a single-ended signal can be deve loped at the secondary windi ng by grounding one side of the output winding. this method will of ten give better dist ortion performance at high frequencies than simply taking the output si gnal directly from one of the dac current outputs and grounding the other. modern current output dacs usually have differ ential outputs, to achieve high common-mode rejection and reduce the even-order distortion products. fullscal e output currents in the range of 2 ma to 30 ma are common. in many applications, it is desirable to convert the differential output of the dac into a single- ended signal, suitable for driving a coax line. this can be readily achieved with an rf transformer, provided low frequency response is not required. figure 2 shows a typical example of this approach. the high impedance current output of the dac is terminated differentially with 50 , which defines the source impeda nce to the transformer as 50 . the resulting differential voltage drives the primar y of a 1:1 rf transformer, to develop a single- ended voltage at the output of the secondary winding. the output of the 50- lc filter is matched with the 50- load resistor r l , and a final output voltage of 1-vp-p is developed. lc filter mini-circuits adt1-1wt 1:1 r diff = 50 r load = 50 v load = 0.5v i out i out 0 to 20ma 20 to 0ma 10ma cmos dac figure 2: differential transformer coupling the transformer not only serves to convert the differential output into a single-ended signal, but it also isolates the output of the dac from the r eactive load presented by the lc filter, thereby improving overall distortion performance. an op amp connected as a differe ntial to single-ended converter can be used to obtain a single- ended output when frequency response to dc is required. in figure 3 the ad8055 op amp is used to achieve high bandwidth and low distortion. the current output dac drives balanced 25- resistive loads, thereby developing an out-of-pha se voltage of 0 to +0.5 v at each output. this technique is used in lieu of a direct i/v conversion to prevent fast slewing dac currents from overloading the amplifier and intr oducing distortion. care must be taken so that the dac output voltage is within its compliance rating. page 4 of 14
MT-019 the ad8055 is configured for a gain of 2, to develop a final single-ended ground-referenced output voltage of 2-v p-p. note that because th e output signal swings above and below ground, a dual-supply op amp is required. i out i out 0 to 20ma 20 to 0ma cmos dac ad8055 + ? +5v ?5v 25 25 0v to +0.5v +0.5v to 0v c filter 500 500 1k 1k 1v f 3db = 1 2 ?50 ?c filter figure 3: differential dc coupled ou tput using a dual supply op amp the c filter capacitor forms a differential filter with the equivalent 50- differential output impedance. this filter reduces any slew-induced distortion of the op amp, and the optimum cutoff frequency of the filter is determined em pirically to give the best overall distortion performance. a modified form of the figure 3 circuit can be operated on a singl e supply, provided the common-mode voltage of the op amp is set to mi d-supply (+2.5 v). this is shown in figure 4, where the ad8061 op amp is used. the output voltage is 2-vp-p centered around a common- mode voltage of +2.5 v. this common-mode volta ge can be either deve loped from the +5 v supply using a resistor divider, or directly from a +2.5 v voltage reference. if the +5 v supply is used as the common-mode voltage, it must be heavily decoupled to prevent supply noise from being amplified. page 5 of 14
MT-019 i out i out 0 to 20ma 20 to 0ma cmos dac ad8061 + ? +5v 25 25 0v to +0.5v +0.5v to 0v c filter 500 500 2k 1k 1v f 3db = 1 2 ?50 ?c filter +2.5v +5v 2k +2.5v ref 1k see text +5v figure 4: differential dc coupled out put using a single- supply op amp single-ended current-to- voltage conversion single-ended current-to-voltage conversion is eas ily performed using a single op amp as an i/v converter, as shown in figure 5. the 10- ma full scale dac current from the ad768 develops a 0 to +2 v output voltage across the 200- r f resistor. i out i out 0 to 10ma ad768 16-bit bicmos dac ad8055 + ? +5v r f = 200 ?5v r dac ||c dac c f 0 to +2.0v for r dac r f , make c f c in r dac (c dac + c in ) r f for r dac >> r f , make c f c dac + c in 2 r f fu fu = op amp unity gain-bandwidth product figure 5: single-ended i/v op amp interface for precision 16-bit ad768 dac page 6 of 14
MT-019 driving the virtual ground of the ad8055 op amp minimizes any distortion due to nonlinearity in the dac output impedance. in fact, most high resolution dacs of this type are factory trimmed using an i/v converter. it should be recalled, however, that using the si ngle-ended output of the dac in this manner will cause degradation in the common- mode rejection and increased s econd-order distortion products, compared to a differential operating mode. the c f feedback capacitor should be optimized for best pulse response in the circuit. the equations given in the diagram should only be used as guidelines. an r-2r based current-output dac has a code-dependent output impedance?therefore, its output must drive the virtual ground of an op amp in order to maintain linearity. the ad5545/ad5555 16-/14-bit dac is an excellent exam ple of this architecture. a suitable interface circuit is shown in figure 6 where the adr03 is used as a 2.5-v voltage reference, and the ad8628 chopper-stabilized op amp is used as an output i/v converter. c f i out = 0 to +0.5ma v out = 0 to ?2.5v c f i out = 0 to +0.5ma v out = 0 to ?2.5v figure 6: ad5545/ad5555 dual 16-/14-bit r-2r current output dac interface the external 2.5-v references determines the fulls cale output current, 0 .5 ma. note that a 5-k feedback resistor is included in the dac, and using it will enhance temperature stability as opposed to using an external resistor. the fullsca le output voltage from the op amp is therefore ? 2.5 v. the c f feedback capacitor compensates for th e dac output capacitance and should be selected to optimize the pulse response , with 20 pf a typical starting point. page 7 of 14
MT-019 differential current-to-differential voltage conversion if a buffered differential volta ge output is required from a current output dac, the ad813x- series of differential amplifiers can be used as shown in figure 7. i out i out 0 to 20ma 0 to +0.5v 20 to 0ma +0.5 to 0v cmos dac + ? ad8138 v ocm 2.49k 2.49k 5v p-p differential output 25 25 499 499 figure 7: buffering high speed dacs using the ad8138 differential amplifier the dac output current is first converted into a voltage that is developed across the 25- resistors. the voltage is amp lified by a factor of 5 using the ad8138 . this technique is used in lieu of a direct i/v conversion to prev ent fast slewing dac curre nts from overloading the amplifier and introducing distortion. care must be taken so that the dac out put voltage is within its compliance rating. the v ocm input on the ad8138 can be used to set a final output common-mode voltage within the range of the ad8138. adding a pair of 75- series output resistors will allow transmission lines to be driven. dac data input considerations the earliest monolithic dacs containe d little, if any, logic circuitry, and parallel data had to be maintained on the digital input to maintain the digital output. today almost all dacs are latched, and data need only be written to them, not ma intained. some even have nonvolatile latches and remember settings while turned off. there are innumerable variations of dac input structure, which wi ll not be discussed here, but nearly all are described as "double-buffered." a double-buffered dac has two sets of latches. data is initially latched in the first rank and s ubsequently transferred to the second as shown in figure 8. there are several reasons why this arrangement is useful. page 8 of 14
MT-019 output digital input input structure: may be serial, parallel, byte-wide, etc. output latch transfers data to dac - timing is independent of input dac output strobe - may go to many dacs f c = sampling frequency figure 8: double-buffere d dac permits complex input structures and simultaneous update the first is that it allows data to enter the dac in many different ways. a dac without a latch, or with a single latch, must be loaded in parallel with all bits at once, since otherwise its output during loading may be totally different from wh at it was, or what it is to become. a double- buffered dac, on the other hand, may be loaded with parallel data, or with serial data, with 4-bit or 8-bit words, or whatever, and the output will be unaffected until the new data is completely loaded and the dac receives its update instruction. a second advantage of a double-buffered dac is that the time skew be tween the individual switches is minimized by driving all the switches in parallel with a single latch which is updated at the dac output data rate. this minimizes the glitch impulse and improves distortion performance. the third convenience of the double-buffered st ructure is that many dacs may be updated simultaneously. data is loaded into the first rank of each dac in turn, and when all is ready, the output buffers of all the dacs are updated at on ce. there are many dac applications where the output of a number of dacs mu st change simultaneously, a nd the double-buffered structure allows this to be done very easily. most early monolithic high resolution dacs had parallel or byte-wide data ports and tended to be connected to parallel data bus es and address decode rs and addressed by mi croprocessors as if they were very small write-only memories. (some parallel dacs are not write-only, but can have their contents read as well?this is conv enient for some applications, but is not very common.) a dac connected to a da ta bus is vulnerable to capaci tive coupling of logic noise from the bus to the analog output, and therefore many dacs today have serial data structures. these are less vulnerable to such noise (since fewe r noisy pins are involved), use fewer pins and therefore take less board space, and are freq uently more convenient for use with modern microprocessors, most of which have serial data ports. some, but not all, of such serial dacs have both data outputs a nd data inputs so that several dacs may be connected in series, with page 9 of 14
MT-019 data clocked to all of them from a single serial port. this arrangement is often referred to as "daisy-chaining." serial dacs can be used at voiceband and audio frequency update rates. for instance, 24-bit digital audio updated at 192 ksps re quires a serial port transfer ra te of at least 24 192 ksps = 46.08 msps, which is easily handled by cmos l ogic. however where high update rates are involved, parallel dacs must be used since the required transfer rate of the serial data would be too high. for parallel data rates greater than approxima tely 100 msps, low-level current-mode differential logic (pecl, reduced-level pecl, lvds, etc.) is often used because it is much less likely to generate transient glitches than cmos logic leve ls (see figure 9). this helps minimize distortion generated by code-dependent glitches. for instance, the ad9734/ad9735/ad9736 dac family operates at 1.2 gsps and accepts lvds input logic levels. special circuitry is included on-chip to ensure the proper timing of the inpu t data with respect to the dac clock. (3.5ma) (3.5ma) +3.3v) 3.5k 3.5k +1.2v (3.5ma) (3.5ma) output driver +3.3v) 3.5k 3.5k +1.2v 3.5k 3.5k +1.2v v+ v+ v? v? figure 9: lvds driver dac clock considerations it was shown in tutorial mt-007 that the relationship between adc broadband aperture jitter, t j , converter snr, and fullscale sinewave analog input frequency, f, is given by ? ? ? ? ? ? ? ? = j 10 tf2 1 log20 snr . eq. 1 page 10 of 14
MT-019 the same relationship is applicable to rec onstruction dacs. the equation assumes an ideal adc/dac, where the only error source is clock jitter. the bandwidth for the snr measurement is the nyquist bandwidth, dc to f c /2, where f c is the dac update rate. note that eq. 1 also assumes a fullscale sinewave output. the error due to jitter is proportional to the slew rate of the output signal?lower amplitude sinewaves with proportionally lower slew rate yield higher values of snr (with respect to fullscale). it should be noted that t j in eq. 1 is the combined jitter of the sampling clock, t jc , and the adc internal aperture jitter, t ja ?these terms are not correlated and therefore combine on an root-sum- square (rss) basis: 2 ja 2 jcj ttt += . eq. 2 high-speed reconstruction dacs, on the other ha nd, do not have specifica tions for internal aperture jitter because they have no internal sample-and-hold amplifier. although there is an internal clock jitter component in a dac, it is generally not meas ured or specified, because the jitter of the external clock is the dominant jitter source. snr (db) enob 100 80 60 40 20 16 14 12 10 8 6 4 13 10 30 100 t j = 1ns t j = 100ps t j = 10ps t j = 1ps t j = 0.1ps 120 18 full-scale sinewave analog input frequency (mhz) snr = 20log 10 1 2 ft j snr = 20log 10 1 2 ft j t j = 50fs figure 10: theoretical snr and enob due to jitter vs. fullscale sinewave analog output frequency figure 10 plots eq. 1 and graphically illustrate s how snr is degraded by jitter for various fullscale analog output frequencie s (note that we assume t j includes all jitter sources, including the internal dac jitter). for instance, maintain ing 12-bit snr (74 db) for a 70 mhz if output frequency requires the clock jitter to be less than 0.45 ps (using eq. 1). page 11 of 14
MT-019 from tutorial mt-001 , it was shown that there is a very useful relationship between effective number of bits (enob) and the signal-to-noi se-plus-distortion ratio (sinad) given by: db02.6 db76.1sinad enob ? = . eq. 3 for the purposes of this discussion, assume that the dac has no distortion, and therefore sinad = snr, so eq. 3 becomes: db02.6 db76.1snr enob ? = . eq. 4 the snr values on the left-hand vertical axis of figure 10 have been converted into enob values on the right-hand vert ical axis using eq. 4. in order to illustrate the significance of these jitter numbers, consider the typical rms jitter associated with a selection of logic gates sh own in figure 11. the values for the 74ls00, 74hct00, and 74act00 were measured with a hi gh performance adc (aperture jitter less than 0.2-ps rms) using the method described in ch apter 5 of reference 1, where the jitter was calculated from fft-based snr degradation due to several identical gates connected in series. the jitter due to a single gate was then calcul ated by dividing by the squa re root of the total number of series-connected gates. the jitte r for the mc100el16 and nbsg16 was specified by the manufacturer. ? 74ls00 4.94 ps * ? 74hct00 2.20 ps * ? 74act00 0.99 ps * ? mc100el16 pecl 0.7 ps ** ? nbsg16, reduced swing ecl (0.4v) 0.2 ps ** z * calculated values based on degradation in adc snr z ** manufacturers' specification figure 11: rms jitter of typical logic gates figure 12 shows the same data as figure 10 but pl ots maximum allowable ji tter as a function of analog output frequency for various resolution re quirements. this graph should serve as an approximate guideline for selecting the type of sampling clock generator based upon the maximum output frequency and the required re solution in enob. the pll approach with a standard vco is an excellent one for generating sampling clocks where the rms jitter requirement is approximately 1 ps or greater. ho wever, sub-picosecond jitter requires either a page 12 of 14
MT-019 vcxo-based pll or a dedicated low noise crysta l oscillator. tutorial mt-008 explains how to convert oscillator phase noise into jitter. 1 10 100 1000 0.1 1 10 100 1000 33 03 0 0 0.3 3 30 300 full-scale analog output frequency (mhz) t j (ps) 16 14 12 10 8 6 0.1 0.3 1 3 10 30 100 300 1000 18 4 t j (ps) enob = snr ?1.76db 6.02 0.03 0.03 pll with vco pll with vcxo dedicated low noise xtal osc figure 12: oscillator require ments vs. resolution and analog output frequency this section has described the effects of jitter on snr, assuming that the jitter is solely a combination of the internal dac jitter and the external clock jitter. however, improper layout, grounding, and decoupling techniques can create additional clock jitter which ca n drastically degrade dynamic performance, regardless of th e specifications of th e dac or sampling clock oscillator. routing the sampling clock signal in parallel with noisy digital signals is sure to degrade performance due to stray coupling. in fact, c oupling high speed data from parallel output adcs into the sampling clock not only in creases noise, but is likely to create additional harmonic distortion, because the energy contained in th e digital output transient currents is signal dependent. for further discussion of these and ot her critical hardware design techniques, the reader is referred to chapter 9 of reference 1. page 13 of 14
page 14 of 14 MT-019 reference 1. walt kester, analog-digital conversion , analog devices, 2004, isbn 0-916550-27-3. also available as the data conversion handbook , elsevier/newnes, 2005, isbn 0-7506-7841-0. copyright 2009, analog devices, inc. all rights reserved. analog devices assumes no responsibility for customer product design or the use or application of customers? products or for any infringements of patents or rights of others which may result from analog devices assistance. all trad emarks and logos are property of their respective holders. information furnished by analog devices applications and development tools engineers is believed to be accurate and reliable, however no responsibility is assumed by analog devices regarding technical accuracy and topicality of the content provided in analog devices tutorials.

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