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general description the MAX1953/max1954/max1957 is a family of versa- tile, economical, synchronous current-mode, pulse-width modulation (pwm) buck controllers. these step-down controllers are targeted for applications where cost and size are critical. the MAX1953 operates at a fixed 1mhz switching fre- quency, thus significantly reducing external component size and cost. additionally, excellent transient response is obtained using less output capacitance. the MAX1953 operates from low 3v to 5.5v input voltage and can sup- ply up to 10a of output current. selectable current limit is provided to tailor to the external mosfets?on-resistance for optimum cost and performance. the output voltage is adjustable from 0.8v to 0.86v in . with the max1954, the drain-voltage range on the high- side fet is 3v to 13.2v and is independent of the supply voltage. it operates at a fixed 300khz switching frequen- cy and can be used to provide up to 25a of output cur- rent with high efficiency. the output voltage is adjustable from 0.8v to 0.86v hsd . the max1957 features a tracking output voltage range of 0.4v to 0.86v in and is capable of sourcing or sinking current for applications such as ddr bus termination and powerpc/asic/dsp core supplies. the max1957 operates from a 3v to 5.5v input voltage and at a fixed 300khz switching frequency. the MAX1953/max1954/max1957 provide a comp pin that can be pulled low to shut down the converter in addition to providing compensation to the error amplifier. an input undervoltage lockout (ulvo) is provided to ensure proper operation under power-sag conditions to prevent the external power mosfets from overheating. internal digital soft-start is included to reduce inrush cur- rent. the MAX1953/max1954/max1957 are available in tiny 10-pin ?ax packages. applications printers and scanners graphic cards and video cards pcs and servers microprocessor core supply low-voltage distributed power telecommunications and networking features low-cost current-mode controllers fixed-frequency pwm MAX1953 1mhz switching frequency small component size, low cost adjustable current limit max1954 3v to 13.2v input voltage 25a output current capability 93% efficiency 300khz switching frequency max1957 tracking 0.4v to 0.86v in output voltage range sinking and sourcing capability of 3a shutdown feature all n-channel mosfet design for low cost no current-sense resistor needed internal digital ? soft-start thermal overload protection small 10-pin max package MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller ________________________________________________________________ maxim integrated products 1 ordering information 19-2373; rev 0; 4/02 for pricing, delivery, and ordering information, please contact maxim/dallas direct! at 1-888-629-4642, or visit maxim? website at www.maxim-ic.com. part temp range pin-package MAX1953 eub -40 c to +85 c 10 max max1954 eub -40 c to +85 c 10 max max1957 eub -40 c to +85 c 10 max top view 1 2 3 4 5 10 9 8 7 6 bst lx dh pgnd gnd fb comp ilim MAX1953eub max dl in pin configurations pin configurations continued at end of data sheet. ? patent pending powerpc is a trademark of motorola, inc.
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller 2 _______________________________________________________________________________________ absolute maximum ratings stresses beyond those listed under ?bsolute maximum ratings?may cause permanent damage to the device. these are stress rating s only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specificatio ns is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. in, fb to gnd...........................................................-0.3v to +6v lx to bst..................................................................-6v to +0.3v bst to gnd ............................................................-0.3v to +20v dh to lx ....................................................-0.3v to (v bst + 0.3v) dl, comp to gnd.......................................-0.3v to (v in + 0.3v) hsd, ilim, refin to gnd ........................................-0.3v to 14v pgnd to gnd .......................................................-0.3v to +0.3v i dh , i dl ................................................................100ma (rms) continuous power dissipation (t a = +70 c) (derate 5.6mw/ c above +70 c) ..................................444mw operating temperature range ...........................-40 c to +85 c junction temperature ......................................................+150 c storage temperature range .............................-65 c to +150 c lead temperature (soldering, 10s) .................................+300 c electrical characteristics (v in = 5v, v bst - v lx = 5v, t a = -40 c to +85 c, unless otherwise noted. typical values are at t a = +25 c.) (note 1) parameter conditions min typ max units operating input voltage range 3.0 5.5 v hsd voltage range max1954 only (note 2) 3.0 13.2 v quiescent supply current v fb = 1.5v, no switching 1 2 ma standby supply current (MAX1953/ max1957) v in = v bst = 5.5v, comp = gnd 220 350 a standby supply current (max1954) v in = v bst = 5.5v, v hsd = 13.2v, comp = gnd 220 350 a undervoltage lockout trip level rising and falling v in , 3% hysteresis 2.50 2.78 2.95 v output voltage adjust range (v out ) 0.8 0.86 x v in v error amplifier t a = 0 c to +85 c (MAX1953/max1954) 0.788 0.8 0.812 t a = -40 c to +85 c (MAX1953/max1954) 0.776 0.8 0.812 fb regulation voltage max1957 only v refin - 8mv v refin v refin + 8mv v transconductance 70 110 160 s fb input leakage current v fb = 0.9v 5 500 na refin input bias current v refin = 0.8v, max1957 only 5 500 na fb input common-mode range -0.1 1.5 v refin input common-mode range max1957 only -0.1 1.5 v current-sense amplifier voltage gain low ilim = gnd (MAX1953 only) 5.67 6.3 6.93 v/v v ilim = v in or ilim = open (MAX1953 only) current-sense amplifier voltage gain max1954/max1957 3.15 3.5 3.85 v/v
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller _______________________________________________________________________________________ 3 parameter conditions min typ max units ilim input impedance MAX1953 only 50 125 200 k ? v pgnd - v lx , ilim = gnd (MAX1953 only) 85 105 125 v pgnd - v lx , ilim = open (MAX1953 only) 190 210 235 v pgnd - v lx , ilim = in (MAX1953 only) 290 320 350 current-limit threshold v pgnd v lx (max1954/max1957 only) 190 210 235 mv oscillator MAX1953 0.8 1 1.2 mhz switching frequency max1954/max1957 240 300 360 khz maximum duty cycle measured at dh 86 89 96 % MAX1953, measured at dh 15 18 minimum duty cycle max1954/max1957, measured at dh 4.5 5.5 % soft-start MAX1953 4 soft-start period max1954/max1957 3.4 ms fet drivers dh on-resistance, high state 2 3 ? dh on-resistance, low state 1.5 3 ? dl on-resistance, high state 2 3 ? dl on-resistance, low state 0.8 2 ? lx, bst leakage current v bst = 10.5v, v lx = v in = 5.5v, MAX1953/max1957 20 a lx, bst, hsd leakage current v bst = 18.7v, v lx = 13.2v, v in = 5.5v v hsd = 13.2v (max1954 only) 30 a thermal protection thermal shutdown rising temperature 160 c thermal shutdown hysteresis 15 c shutdown control comp logic level low 3v < v in < 5.5v 0.25 v comp logic level high 3v < v in < 5.5v 0.8 v comp pullup current 100 a electrical characteristics (continued) (v in = 5v, v bst - v lx = 5v, t a = -40 c to +85 c, unless otherwise noted. typical values are at t a = +25 c.) (note 1) note 1: specifications to -40 c are guaranteed by design and not production tested. note 2: hsd and in are externally connected for applications where v hsd < 5.5v.
typical operating characteristics (t a = +25 c, unless otherwise noted.) MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller 4 _______________________________________________________________________________________ max1954 efficiency vs. load current MAX1953 toc02 load current (a) efficiency (%) 1 50 60 70 80 90 100 40 0.1 10 v out = 2.5v v out = 1.7v v in = 5v circuit of figure 2 MAX1953 output voltage vs. load current MAX1953 toc05 load current (a) output voltage (v) 2.5 2.0 1.5 1.0 0.5 2.45 2.50 2.55 2.60 2.40 0 3.0 v in = 5v v in = 3.3v circuit of figure 1 max1954 output voltage vs. load current MAX1953 toc06 load current (a) output voltage (v) 5 4 3 2 1 2.40 2.45 2.50 2.55 2.60 2.35 06 v hsd = v in = 5v circuit of figure 2 MAX1953 efficiency vs. load current MAX1953 toc01 load current (a) efficiency (%) 1 55 60 65 70 75 80 85 90 95 100 50 0.1 10 v in = 3.3v v in = 5v v out = 2.5v circuit of figure 1 max1957 efficiency vs. load current MAX1953 toc03 load current (a) efficiency (%) 1 50 60 70 80 90 100 40 0.1 10 v out = 1.25v v in = 5v circuit of figure 3 max1954 efficiency vs. load current MAX1953 toc04 load current (a) efficiency (%) 20 15 10 5 55 60 65 70 75 80 85 90 95 100 50 025 v out = 1.8v v in = 12v circuit of figure 4 max1954 output voltage vs. load current MAX1953 toc07 load current (a) output voltage (v) 5 4 3 2 1 1.60 1.65 1.70 1.75 1.80 1.55 06 v hsd = v in = 5v circuit of figure 2
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller _______________________________________________________________________________________ 5 typical operating characteristics (continued) (t a = +25 c, unless otherwise noted.) MAX1953 frequency vs. input voltage MAX1953 toc13 input voltage (v) frequency (mhz) 5.0 4.5 4.0 3.5 0.98 1.00 1.02 1.04 1.06 0.96 3.0 5.5 t a = -40 c t a = +85 c t a = +25 c v out = 2.5v max1954/max1957 frequency vs. input voltage MAX1953 toc14 input voltage (v) frequency (khz) 5.0 4.5 4.0 3.5 275 280 285 290 295 300 305 310 315 320 270 3.0 5.5 t a = -40 c t a = +25 c t a = +85 c v out = 1.25v max1957 output voltage vs. load current MAX1953 toc08 load current (a) output voltage (v) 2 1 0 -1 -2 1.20 1.25 1.30 1.35 1.15 -3 3 v in = 5v circuit of figure 3 max1954 output voltage vs. input voltage MAX1953 toc11 input voltage (v) output voltage (v) 5.0 4.5 4.0 3.5 2.47 2.48 2.49 2.50 2.51 2.52 2.46 3.0 5.5 i load = 0 i load = 5a circuit of figure 2 max1957 output voltage vs. input voltage MAX1953 toc12 input voltage (v) output voltage (v) 5.0 4.5 4.0 3.5 1.21 1.23 1.25 1.27 1.29 1.19 3.0 5.5 i load = 0 i load = 3a circuit of figure 3 MAX1953 output voltage vs. input voltage MAX1953 toc09 input voltage (v) output voltage (v) 5.0 4.5 4.0 3.5 2.45 2.50 2.55 2.60 2.40 3.0 5.5 i load = 3a i load = 0 circuit of figure 1 max1954 output voltage vs. input voltage MAX1953 toc10 input voltage (v) output voltage (v) 5.0 4.5 4.0 3.5 1.66 1.68 1.70 1.72 1.74 1.76 1.64 3.0 5.5 i load = 0 i load = 5a circuit of figure 2
typical operating characteristics (continued) (t a = +25 c, unless otherwise noted.) MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller 6 _______________________________________________________________________________________ MAX1953 no-load switching waveforms MAX1953 toc18 2 s/div i lx lx dl dh 2a/div 5v/div 5v/div 5v/div max1957 load transient MAX1953 toc17 v out ac-coupled i load 50mv/div 3a -3a 400 s/div MAX1953 load transient circuit of figure 1 400 s/div 100mv/div 3a 1.5a v out ac-coupled i load MAX1953 toc15 max1954 load transient MAX1953 toc16 100mv/div 2.5a 5a i load v out ac-coupled 400 s/div
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller _______________________________________________________________________________________ 7 typical operating characteristics (continued) (t a = +25 c, unless otherwise noted.) MAX1953 full-load switching waveforms MAX1953 toc19 2 s/div i lx lx dl dh 2a/div 5v/div 5v/div 5v/div max1954/max1957 short-circuit switching waveforms MAX1953 toc23 4 s/div i lx lx dl dh 5a/div 10v/div 5v/div 10v/div max1954/max1957 full-load switching waveforms MAX1953 toc22 4 s/div i lx lx dl dh 2a/div 10v/div 5v/div 10v/div max1954/max1957 no-load switching waveforms MAX1953 toc21 4 s/div i lx lx dl dh 2a/div 10v/div 5v/div 10v/div MAX1953 short-circuit switching waveforms MAX1953 toc20 2 s/div i lx lx dl dh 5a/div 5v/div 5v/div 5v/div
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller 8 _______________________________________________________________________________________ pin description pin MAX1953 max1954 max1957 name function 1 ilim ilim sets the current-limit threshold for the low-side n-channel mosfet, as well as the current-sense amplifier gain. connect to in for 320mv, leave floating for 210mv, or connect to gnd for 105mv current-limit threshold. 1 hsd hsd senses the voltage at the drain of the high-side n-channel mosfet. connect to the high-side mosfet drain using a kelvin connection. 1 refin refin sets the fb regulation voltage. drive refin with the desired fb regulation voltage using an external resistor-divider. bypass to gnd with a 0.1f capacitor. 2 2 2 comp compensation and shutdown control pin. connect an rc network to compensate control loop. drive to gnd to shut down the ic. 3 3 3 fb feedback input. regulates at v fb = 0.8v (MAX1953/max1954) or refin (max1957). connect fb to a resistor-divider to set the output voltage (MAX1953/max1954). connect to output through a decoupling resistor (max1957). 4 4 4 gnd ground 5 5 5 in input voltage (3v to 5.5v). provides power for the ic. for the MAX1953/max1957, in serves as the current-sense input for the high- side mosfet. connect to the drain of the high-side mosfet (MAX1953/max1957). bypass in to gnd close to the ic with a 0.22f (max1954) capacitor. bypass in to gnd close to the ic with 10f and 4.7f in parallel (MAX1953/max1957) capacitors. use ceramic capacitors. 6 6 6 dl low - s i d e g ate- d r i ve outp ut. d r i ves the synchr onous- r ecti fi er m o s fe t. s w i ng s fr om p gn d to v in . 7 7 7 pgnd power ground. connect to source of the synchronous rectifier close to the ic. 8 8 8 dh high-side gate-drive output. drives the high-side mosfet. dh is a floating driver output that swings from v lx to v bst . 9 9 9 lx master controller current-sense input. connect lx to the junction of the mosfets and inductor. lx is the reference point for the current limit. 10 10 10 bst boost capacitor connection for high-side gate driver. connect a 0.1f ceramic capacitor from bst to lx and a schottky diode to in.
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller _______________________________________________________________________________________ 9 reference and soft-start dac refin (max1957 only) gnd fb comp 0.5v shutdown comparator error amplifier slope compensation thermal limit uvlo pwm control circuitry clock short-circuit current-limit circuitry ilim (MAX1953 only) current-limit comparator pgnd dl in lx current- sense circuitry dh bst hsd (max1954 only) in MAX1953 max1954 max1957 functional diagram ilim comp gnd bst dh lx dl pgnd fb in MAX1953 output 0.8v to 0.86v in input 3v to 5.5v typical operating circuit
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller 10 ______________________________________________________________________________________ detailed description the MAX1953/max1954/max1957 are single-output, fixed-frequency, current-mode, step-down, pwm, dc- dc converter controllers. the MAX1953 switches at 1mhz, allowing the use of small external components for small applications. table 1 lists suggested components. the max1954 switches at 300khz for higher efficiency and operates from a wider range of input voltages. figure 1 is the MAX1953 typical application circuit. the MAX1953/max1954/max1957 are designed to drive a pair of external n-channel power mosfets in a syn- chronous buck topology to improve efficiency and cost compared with a p-channel power mosfet topology. the on-resistance of the low-side mosfet is used for short-circuit current-limit sensing, while the high-side mosfet on-resistance is used for current-mode feed- back and current-limit sensing, thus eliminating the need for current-sense resistors. the MAX1953 has three selectable short-circuit current-limit thresholds: 105mv, 210mv, and 320mv. the max1954 and max1957 have 210mv fixed short-circuit current-limit thresholds. the MAX1953/max1954/max1957 accept input voltages from 3v to 5.5v. the max1954 is config- ured with a high-side drain input (hsd) allowing an extended input voltage range of 3v to 13.2v that is independent of the input supply (figure 2). the max1957 is tailored for tracking output voltage applica- tions such as ddr bus termination supplies, referred to as v tt . it utilizes a resistor-divider network connected to refin to keep the 1/2 ratio tracking between v tt and v ddq (figure 3). the max1957 can source and sink up to 3a. figure 4 shows the max1954 20a circuit. dc-dc converter control architecture the MAX1953/max1954/max1957 step-down convert- ers use a pwm, current-mode control scheme. an inter- nal transconductance amplifier establishes an integrated error voltage. the heart of the pwm controller is an open- loop comparator that compares the integrated voltage- feedback signal against the amplified current-sense signal plus the slope compensation ramp, which are summed into the main pwm comparator to preserve inner-loop stability and eliminate inductor staircasing. at each rising edge of the internal clock, the high-side mosfet turns on until the pwm comparator trips or the maximum duty cycle is reached. during this on-time, cur- rent ramps up through the inductor, storing energy in a magnetic field and sourcing current to the output. the current-mode feedback system regulates the peak inductor current as a function of the output voltage error signal. the circuit acts as a switch-mode transconduc- tance amplifier and pushes the output lc filter pole nor- mally found in a voltage-mode pwm to a higher frequency. during the second half of the cycle, the high-side mos- fet turns off and the low-side mosfet turns on. the inductor releases the stored energy as the current ramps down, providing current to the output. the output capaci- tor stores charge when the inductor current exceeds the required load current and discharges when the inductor current is lower, smoothing the voltage across the load. under overload conditions, when the inductor current exceeds the selected current-limit (see the current limit circuit section), the high-side mosfet is not turned on at the rising clock edge and the low-side mosfet remains on to let the inductor current ramp down. the MAX1953/max1954/max1957 operate in a forced- pwm mode. as a result, the controller maintains a con- stant switching frequency, regardless of load, to allow for easier postfiltering of the switching noise. designation MAX1953 max1954 max1957 20a circuit c1 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg 0.22f, 10v x7r cer kemet c0603c224m8rac 3 x 22f, 6.3v x5r cer taiyo yuden jmk316bj226ml 0.22f, 10v x7r cer kemet c0603c224m8rac c2 0.1f, 50v x7r cer taiyo yuden umk107bj104ka 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg 0.1f, 50v x7r cer taiyo yuden umk107bj104ka 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg c3 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg 0.1f, 50v x7r cer taiyo yuden umk107bj104ka 270f, 2v sp polymer panasonic eefueod271r 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg table 1. suggested components
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller ______________________________________________________________________________________ 11 designation MAX1953 max1954 max1957 20a circuit c4 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg 180f, 4v sp polymer panasonic eefueog181r 270f, 2v sp polymer panasonic eefueod271r 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg c5 4.7f, 6.3v x5r cer taiyo yuden jmk212bj475mg 270f, 2v sp polymer panasonic eefueod271r 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg c6 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg 10f, 6.3v x5r cer taiyo yuden jmk212bj106mg c7 4.7f, 6.3v x5r cer taiyo yuden jmk212bj475mg 0.1f, 50v x7r cer taiyo yuden umk107bj104ka c8 0.1f, 50v x7r cer taiyo yuden umk107bj104ka 270f, 2v sp polymer panasonic eefueod271r c9-c13 270f, 2v sp polymer panasonic eefueod271r c14 1500pf, 50v x7r cer murata grm39x7r152k50 c c 270pf, 10v x7r cer kemet c0402c271m8rac 1000pf, 10v x7r cer kemet c0402c102m8rac 470pf, 50v x7r cer murata grm39x7r471k50 560pf, 10v x7r cer kemet c0402c561m8rac c f 47pf, 10v c0g cer kemet c0402c470k8gac 68pf, 50v cog cer murata grm39cog680j50 15pf, 10v c0g cer kemet c0402c150k8gac d1 schottky diode central semiconductor cmpsh1-4 schottky diode central semiconductor cmpsh1-4 schottky diode central semiconductor cmpsh1-4 schottky diode central semiconductor cmpsh1-4 l1 1h 3.6a toko 817fy-1r0m 2.7h 6.6a coilcraft do3316-272hc 2.7h 6.6a coilcraft do3316-272hc 0.8h 27.5a sumida cep125u-0r8 n1-n2 dual mosfet 20v 5a fairchild fds6898a dual mosfet 20v fairchild fds6890a dual mosfet 20v fairchild fds6898a n-channel 30v international rectifier irf7811w n3-n4 n-channel 30v siliconix si4842dy r1 16.9k ? 1% 9.09k ? 1% 2k ? 1% 10k ? 1% r2 8.06k ? 1% 8.06k ? 1% 2k ? 1% 8.06k ? 1% r3 10k ? 5% r c 33k ? 5% 62k ? 5% 51.1k ? 5% 270k ? 5% table 1. suggested components (continued)
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller 12 ______________________________________________________________________________________ ilim comp gnd bst dh lx dl pgnd fb in MAX1953 v out 2.5v at 3a v in 3v to 5.5v n1 d1 c5 4.7 f c6 10 f c1 10 f c2 0.1 f c3 10 f c4 10 f l1 1 h c c 270pf r c 33k ? r1 16.9k ? r2 8.06 ? figure 1. typical application circuit for the MAX1953 hsd comp gnd bst dh lx dl pgnd fb in max1954 v out 1.7v at 3a v in 3v to 5.5v n1 d1 v hsd 5.5v to 13.2v c2 10 f l1 2.7 h c3 0.1 f c4 180 f r1 9.09k ? r2 8.06k ? c1 0.22 f c c 1000pf c f 47pf r c 62k ? figure 2. typical application circuit for the max1954
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller ______________________________________________________________________________________ 13 figure 3. typical application circuit for the max1957 comp refin gnd bst dh lx dl pgnd fb in max1957 v tt = 1/2 v ddq v in 3v to 5.5v n1 d1 c7 4.7 f c6 10 f c1 3 ? 22 f c2 0.1 f c3 270 f c4 270 f c5 270 f l1 2.7 h r3 10k ? c c 470pf c f 68pf r c 51.1k ? c8 0.1 f r1 2k ? r2 2k ? v ddq c14 1500pf in comp gnd bst dh lx dl pgnd fb hsd max1954 v out 1.8v at 20a v in 3v to 5.5v d1 c6 10 f c5 10 f c4 10 f c3 10 f c2 10 f l1 0.8 h c7 0.1 f c13 270 f c12 270 f c11 270 f c10 270 f c9 270 f c8 270 f r1 10k ? r2 8.06k ? c1 0.22 f c c 560pf c f 15pf r c 270k ? n1 n2 n3 n4 v hsd 10.8v to 13.2v figure 4. 20a circuit
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller 14 ______________________________________________________________________________________ current-sense amplifier the MAX1953/max1954/max1957s current-sense cir- cuit amplifies (a v = 3.5 typ) the current-sense voltage (the high-side mosfet s on-resistance (r ds(on) ) multi- plied by the inductor current). this amplified current- sense signal and the internal-slope compensation signal are summed (v sum ) together and fed into the pwm comparator s inverting input. the pwm compara- tor shuts off the high-side mosfet when v sum exceeds the integrated feedback voltage (v comp ). current-limit circuit the current-limit circuit employs a lossless current-limit- ing algorithm that uses the low-side and high-side mosfets on-resistances as the sensing elements. the voltage across the high-side mosfet is monitored for current-mode feedback, as well as current limit. this signal is amplified by the current-sense amplifier and is compared with a current-sense voltage. if the current- sense signal is larger than the set current-limit voltage, the high-side mosfet turns off. once the high-side mosfet turns off, the low-side mosfet is monitored for current limit. if the voltage across the low-side mos- fet (r ds(on) ? i inductor ) does not exceed the short- circuit current limit, the high-side mosfet turns on normally. in this condition, the output drops smoothly out of regulation. if the voltage across the low-side mosfet exceeds the short-circuit current-limit thresh- old at the beginning of each new oscillator cycle, the MAX1953/max1954/max1957 do not turn on the high- side mosfet. in the case where the output is shorted, the low-side mosfet is monitored for current limit. the low-side mosfet is held on to let the current in the inductor ramp down. once the voltage across the low-side mosfet drops below the short-circuit current-limit threshold, the high-side mosfet is pulsed. under this condition, the frequency of the MAX1953/max1954/ max1957 appears to decrease because the on-time of the low-side mosfet extends beyond a clock cycle. the actual peak output current is greater than the short-circuit current-limit threshold by an amount equal to the inductor ripple current. therefore, the exact cur- rent-limit characteristic and maximum load capability are a function of the low-side mosfet on-resistance, inductor value, input voltage, and output voltage. the short-circuit current-limit threshold is preset for the max1954/max1957 at 210mv. the MAX1953, however, has three options for the current-limit threshold: con- nect ilim to in for a 320mv threshold, connect ilim to gnd for 105mv, or leave floating for 210mv. synchronous rectifier driver (dl) synchronous rectification reduces conduction losses in the rectifier by replacing the normal schottky catch diode with a low-resistance mosfet switch. the MAX1953/max1954/max1957 use the synchronous rectifier to ensure proper startup of the boost gate- driver circuit and to provide the current-limit signal. the dl low-side waveform is always the complement of the dh high-side drive waveform. a dead-time circuit moni- tors the dl output and prevents the high-side mosfet from turning on until dl is fully off, thus preventing cross-conduction or shoot-through. in order for the dead-time circuit to work properly, there must be a low- resistance, low-inductance path from the dl driver to the mosfet gate. otherwise, the sense circuitry in the MAX1953/max1954/max1957 can interpret the mos- fet gate as off when gate charge actually remains. the dead time at the other edge (dh turning off) is determined through gate sensing as well. high-side gate-drive supply (bst) gate-drive voltage for the high-side switch is generated by a flying capacitor boost circuit (figure 5). the capacitor between bst and lx is charged from the v in supply up to v in , minus the diode drop while the low- side mosfet is on. when the low-side mosfet is switched off, the stored voltage of the capacitor is stacked above lx to provide the necessary turn-on voltage (v gs ) for the high-side mosfet. the controller then closes an internal switch between bst and dh to turn the high-side mosfet on. undervoltage lockout if the supply voltage at in drops below 2.75v, the MAX1953/max1954/max1957 assume that the supply voltage is too low to make valid decisions, so the uvlo circuitry inhibits switching and forces the dl and dh gate drivers low. after the voltage at in rises above 2.8v, the controller goes into the startup sequence and resumes normal operation. startup the MAX1953/max1954/max1957 start switching when the voltage at in rises above the uvlo threshold. however, the controller is not enabled unless all four of the following conditions are met: v in exceeds the 2.8v uvlo threshold. the internal reference voltage exceeds 92% of its nominal value (v ref > 1 v). the internal bias circuitry powers up. the thermal overload limit is not exceeded.
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller ______________________________________________________________________________________ 15 once these conditions are met, the step-down controller enables soft-start and starts switching. the soft-start cir- cuitry gradually ramps up to the feedback-regulation voltage in order to control the rate-of-rise of the output voltage and reduce input surge currents during startup. the soft-start period is 1024 clock cycles (1024/f s , max1954/max1957) or 4096 clock cycles (4096/f s , MAX1953) and the internal soft-start dac ramps the voltage up in 64 steps. the output reaches regulation when soft-start is completed, regardless of output capacitance and load. shutdown the MAX1953/max1954/max1957 feature a low-power shutdown mode. use an open-collector transistor to pull comp low to shut down the ic. during shutdown, the output is high impedance. shutdown reduces the quiescent current (i q ) to approximately 220a. thermal overload protection thermal overload protection limits total power dissipation in the MAX1953/max1954/max1957. when the junction temperature exceeds t j = +160 c, an internal thermal sensor shuts down the device, allowing the ic to cool. the thermal sensor turns the ic on again after the junc- tion temperature cools by 15 c, resulting in a pulsed out- put during continuous thermal overload conditions. design procedures setting the output voltage to set the output voltage for the MAX1953/max1954, connect fb to the center of an external resistor-divider connected between the output to gnd (figures 1 and 2). select r2 between 8k ? and 24k ? , and then calcu- late r1 by: where v fb = 0.8v. r1 and r2 should be placed as close to the ic as possible. for the max1957, connect fb directly to the output through a decoupling resistor of 10k ? to 21k ? (figure 3). the output voltage is then equal to the voltage at refin. again, this resistor should be placed as close to the ic as possible. determining the inductor value there are several parameters that must be examined when determining which inductor is to be used. input voltage, output voltage, load current, switching frequen- cy, and lir. lir is the ratio of inductor current ripple to dc load current. a higher lir value allows for a smaller inductor, but results in higher losses and higher output ripple. a good compromise between size, efficiency, and cost is an lir of 30%. once all of the parameters are chosen, the inductor value is determined as follows: where f s is the switching frequency. choose a standard value close to the calculated value. the exact inductor value is not critical and can be adjusted in order to make trade-offs among size, cost, and efficiency. lower inductor values minimize size and cost, but they also increase the output ripple and reduce the efficiency due to higher peak currents. by contrast, higher inductor val- ues increase efficiency, but eventually resistive losses due to extra turns of wire exceed the benefit gained from lower ac current levels. for any area-restricted applications, find a low-core loss inductor having the lowest possible dc resistance. ferrite cores are often the best choice, although pow- dered iron is inexpensive and can work well at 300khz. l vvv v f i lir out in out in s load max = ? () () rr v v out fb 12 1 = ? ? ? ? ? ? ? figure 5. dh boost circuit bst dh lx dl in MAX1953 max1954 max1957
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller 16 ______________________________________________________________________________________ the chosen inductor s saturation current rating must exceed the expected peak inductor current (ipeak). determine ipeak as: setting the current limit the MAX1953/max1954/max1957 use a lossless cur- rent-sense method for current limiting. the voltage drops across the mosfets created by their on-resis- tances are used to sense the inductor current. calculate the current-limit threshold as follows: where a cs is the gain of the current-sense amplifier. a cs is 6.3 for the MAX1953 when ilim is connected to gnd and 3.5 for the max1954/max1957, and for the MAX1953 when ilim is connected to in or floating. the 0.8v is the usable dynamic range of comp (v comp ). initially, the high-side mosfet is monitored. once the voltage drop across the high-side mosfet exceeds v cs , the high-side mosfet is turned off and the low-side mosfet is turned on. the voltage across the low-side mosfet is then monitored. if the voltage across the low- side mosfet exceeds the short-circuit current limit, a short-circuit condition is determined and the low-side mosfet is held on. once the monitored voltage falls below the short-circuit current-limit threshold, the MAX1953/max1954/max1957 switch normally. the short- circuit current-limit threshold is fixed at 210mv for the max1954/ max1957 and is selectable for the MAX1953. when selecting the high-side mosfet, use the follow- ing method to verify that the mosfet s r ds(on) is suffi- ciently low at the operating junction temperature (t j ): the voltage drop across the low-side mosfet at the valley point and at i load(max) is: where r ds(on) is the maximum value at the desired maximum operating junction temperature of the mos- fet. a good general rule is to allow 0.5% additional resistance for each c of mosfet junction temperature rise. the calculated v valley must be less than v cs . for the MAX1953, connect ilim to gnd for a short- circuit current-limit voltage of 105mv, to v in for 320mv or leave ilim floating for 210mv. mosfet selection the MAX1953/max1954/max1957 drive two external, logic-level, n-channel mosfets as the circuit switch elements. the key selection parameters are: on-resistance (r ds(on) ): the lower, the better. maximum drain-to-source voltage (v dss ): should be at least 20% higher than the input supply rail at the high side mosfet s drain. gate charges (q g , q gd , q gs ): the lower, the better. for a 3.3v input application, choose a mosfet with a rated r ds(on) at v gs = 2.5v. for a 5v input application, choose the mosfets with rated r ds(on) at v gs 4.5v. for a good compromise between efficiency and cost, choose the high-side mosfet (n1) that has conduction losses equal to switching loss at the nominal input volt- age and output current. the selected low-side and high- side mosfets (n2 and n1, respectively) must have r ds(on) that satisfies the current-limit setting condition above. for n2, make sure that it does not spuriously turn on due to dv/dt caused by n1 turning on, as this would result in shoot-through current degrading the efficiency. mosfets with a lower q gd /q gs ratio have higher immu- nity to dv/dt. for proper thermal management design, the power dis- sipation must be calculated at the desired maximum operating junction temperature, t j(max) . n1 and n2 have different loss components due to the circuit oper- ation. n2 operates as a zero-voltage switch; therefore, major losses are the channel conduction loss (p n2cc ) and the body diode conduction loss (p n2dc ): where v f is the body diode forward-voltage drop, t dt is the dead time between n1 and n2 switching transi- tions, and f s is the switching frequency. use r at t p v v ir pivtf ds on j max ncc out in load ds on n dc load f dt s () ( ) () () 2 2 2 1 2 =? = vr i lir i valley ds on load max load max = ? ? ? ? ? ? ? () () ( ) () 2 r v ai ds on n cs peak () . 1 08 v v a cs cs = 08 . ii lir i peak load max load max =+ ? ? ? ? ? ? () () 2
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller ______________________________________________________________________________________ 17 n1 operates as a duty-cycle control switch and has the following major losses: the channel conduction loss (p n1cc ), the voltage and current overlapping switching loss (p n1sw ), and the drive loss (p n1dr ). where i gate is the average dh driver output current capability determined by: where r dh is the high-side mosfet driver s on-resis- tance (3 ? max) and r gate is the internal gate resis- tance of the mosfet (~ 2 ? ): where v gs ~ v in . in addition to the losses above, allow about 20% more for additional losses due to mosfet output capacitances and n2 body diode reverse recov- ery charge dissipated in n1 that exists, but is not well defined in the mosfet data sheet. refer to the mos- fet data sheet for the thermal-resistance specification to calculate the pc board area needed to maintain the desired maximum operating junction temperature with the above calculated power dissipations. the minimum load current must exceed the high-side mosfet s maximum leakage current over temperature if fault conditions are expected. input capacitor the input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuit s switching. the input capacitor must meet the ripple current requirement (i rms ) imposed by the switching currents defined by the following equation: i rms has a maximum value when the input voltage equals twice the output voltage (v in = 2 x v out ), where i rms(max) = i load /2. ceramic capacitors are recom- mended due to their low esr and esl at high frequency, with relatively low cost. choose a capacitor that exhibits less than 10 c temperature rise at the maximum operat- ing rms current for optimum long-term reliability. output capacitor the key selection parameters for the output capacitor are the actual capacitance value, the equivalent series resistance (esr), the equivalent series inductance (esl), and the voltage-rating requirements. these para- meters affect the overall stability, output voltage ripple, and transient response. the output ripple has three components: variations in the charge stored in the out- put capacitor, the voltage drop across the capacitor s esr, and the voltage drop across the esl caused by the current into and out of the capacitor: the output voltage ripple as a consequence of the esr, esl, and output capacitance is: where i p-p is the peak-to-peak inductor current (see the determining the inductor value section). these equa- tions are suitable for initial capacitor selection, but final values should be chosen based on a prototype or eval- uation circuit. as a general rule, a smaller current ripple results in less output voltage ripple. since the inductor ripple current is a factor of the inductor value and input voltage, the output voltage ripple decreases with larger inductance, and increases with higher input voltages. ceramic capacitors are recommended for the MAX1953 due to its 1mhz switching frequency. for the max1954/ max1957, using polymer, tantalum, or aluminum elec- trolytic capacitors is recommended. the aluminum electrolytic capacitor is the least expensive; however, it has higher esr. to compensate for this, use a ceramic capacitor in parallel to reduce the switching ripple and noise. for reliable and safe operation, ensure that the capacitor s voltage and ripple-current ratings exceed the calculated values. v i esr v i cf v v l esl i vv fl v v ripple esr pp ripple c pp out s ripple esl in pp in out s out in () () () = ? ? ? ? ? ? = ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? = ? 8 vv v v ripple ripple esr ripple c ripple esl =++ () () ( ) i ivvv v rms load out in out in = ? () pqvf r rr ndr g gs s gate gate dh 1 = + i v rr gate in dh gate ? + 1 2 p v v i r use r at t pvi qq i f ncc out in load ds on ds on j max n sw in load gs gd gate s 1 2 2 = ? ? ? ? ? ? () = + ? ? ? ? ? ? () () ( )
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller 18 ______________________________________________________________________________________ the MAX1953/max1954/max1957s response to a load transient depends on the selected output capacitors. in general, more low-esr output capacitance results in better transient response. after a load transient, the output voltage instantly changes by esr ? ? i load . before the controller can respond, the output voltage deviates further, depending on the inductor and output capacitor values. after a short period of time (see the typical operating characteristics ), the controller responds by regulating the output voltage back to its nominal state. the controller response time depends on its closed-loop bandwidth. with a higher bandwidth, the response time is faster, preventing the output volt- age from further deviation from its regulating value. compensation design the MAX1953/max1954/max1957 use an internal transconductance error amplifier whose output com- pensates the control loop. the external inductor, high- side mosfet, output capacitor, compensation resistor, and compensation capacitors determine the loop sta- bility. the inductor and output capacitors are chosen based on performance, size, and cost. additionally, the compensation resistor and capacitors are selected to optimize control-loop stability. the component values shown in the typical application circuits (figures 1 through 4) yield stable operation over the given range of input-to-output voltages and load currents. the controller uses a current-mode control scheme that regulates the output voltage by forcing the required current through the external inductor. the MAX1953/ max1954/max1957 use the voltage across the high- side mosfet s on-resistance (r ds(on) ) to sense the inductor current. current-mode control eliminates the double pole in the feedback loop caused by the induc- tor and output capacitor, resulting in a smaller phase shift and requiring less elaborate error-amplifier com- pensation. a simple single-series r c and c c is all that is needed to have a stable high bandwidth loop in applications where ceramic capacitors are used for output filtering. for other types of capacitors, due to the higher capacitance and esr, the frequency of the zero created by the capacitance and esr is lower than the desired close loop crossover frequency. another com- pensation capacitor should be added to cancel this esr zero. the basic regulator loop may be thought of as a power modulator, output feedback divider, and an error ampli- fier. the power modulator has dc gain set by g mc x r load , with a pole and zero pair set by r load , the out- put capacitor (c out ), and its equivalent series resis- tance (r esr ). below are equations that define the power modulator: where r load = v out /i out(max) , and g mc = 1/(a cs ? r ds(on) ), where a cs is the gain of the current-sense amplifier and r ds(on) is the on-resistance of the high- side power mosfet. a cs is 6.3 for the MAX1953 when ilim is connected to gnd, and 3.5 for the max1954/ max1957 and for the MAX1953 when ilim is connect- ed to v in or floating. the frequencies at which the pole and zero due to the power modulator occur are deter- mined as follows: the feedback voltage-divider used has a gain of g fb = v fb /v out , where v fb is equal to 0.8v. the transcon- ductance error amplifier has dc gain, g ea(dc) = gm ? r o . r o is typically 10m ? . a dominant pole is set by the compensation capacitor (c c ), the amplifier output resistance (r o ), and the compensation resistor (r c ). a zero is set by the compensation resistor (r c ) and the compensation capacitor (c c ). there is an optional pole set by c f and r c to cancel the output capacitor esr zero if it occurs before crossover frequency (f c ): the crossover frequency (f c ) should be much higher than the power modulator pole f pmod . also, the crossover frequency should be less than 1/5 the switching frequency: ff f pmod c s << < 5 f crr fzea cr fpea cr pdea coc cc fc = + = = 1 2 1 2 1 2 () f c rflr rfl f cr pmod out load s esr load s zmod out esr = () + + () ? ? ? ? ? ? ? ? = 1 2 1 2 gg rfl rfl mod mc load s load s = + () ()
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller ______________________________________________________________________________________ 19 so the loop-gain equation at the crossover frequency is: for the case where f zesr is greater than f c : then r c is calculated as: where g mea = 110s. the error amplifier compensation zero formed by r c and c c should be set at the modulator pole f pmod . c c is calculated by: as the load current decreases, the modulator pole also decreases. however, the modulator gain increases accordingly, and the crossover frequency remains the same. for the case where f zesr is less than f c , add another compensation capacitor c f from comp to gnd to cancel the esr zero at f zesr . c f is calculated by: figure 6 illustrates a numerical example that calculates r c and c c values for the typical application circuit of figure 1 (MAX1953). applications information see table 2 for suggested manufacturers of the com- ponents used with the MAX1953/max1954/max1957. pc board layout guidelines careful pc board layout is critical to achieve low switching losses and clean, stable operation. the switching power stage requires particular attention. follow these guidelines for good pc board layout: 1) place decoupling capacitors as close to ic pins as possible. keep separate power ground plane (con- nected to pin 7) and signal ground plane (connect- ed to pin 4). 2) input and output capacitors are connected to the power ground plane; all other capacitors are con- nected to the signal ground plane. 3) keep the high current paths as short as possible. 4) connect the drain leads of the power mosfet to a large copper area to help cool the device. refer to the power mosfet data sheet for recommended copper area. 5) ensure all feedback connections are short and direct. place the feedback resistors as close to the ic as possible. 6) route high-speed switching nodes away from sensi- tive analog areas (fb, comp). 7) place the high-side mosfet as close as possible to the controller and connect in (MAX1953/max1957) or hsd (max1954) and lx to the mosfet. 8) use very short, wide traces (50mils to 100mils wide if the mosfet is 1in from the device). chip information transistor count: 2930 process: bicmos c rf f c zesr = 1 2 c v i fl v i fl c r c out out max s out out max s out c = + () () () () r v gvg c out mea fb mod f c = () () () () () ggr and gg rfl rfl f f ea f mea c mod f mc load s load s pmod c c c = = + gg v v ea f mod f fb out cc () () = 1 manufacturer component phone website central semiconductor diode 631-435-1110 www.centralsemi.com coilcraft inductors 800-322-2645 www.coilcraft.com fairchild mosfets 800-341-0392 www.fairchildsemi.com kemet capacitors 864-963-6300 www.kemet.com panasonic capacitors 714-373-7366 www.panasonic.com taiyo yuden capacitors 408-573-4150 www.t-yuden.com toko inductors 800-745-8656 www.toko.com table 2. suggested manufacturers
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller 20 ______________________________________________________________________________________ vv ia cf lh r gs aa r g ar s f mhz r v i v a f c rf out out max out esr mea vcs ds on mc vcs ds on s load out out max pmod out load = = = = = = = = = = = === = 25 3 20 1 0 0025 110 63 0 013 1 12 21 1 25 3 0 833 1 2 . . . . . . . () () () () ? ? ? s s load s esr zesr out esr c l rfl rf mhz h mhz h khz f cr f mhz pick the crossover frequency f () () ? ? ? ? ? ? ? ? + = + () + ? ? ? ? ? ? ? ? = = = = ) . . . . . . () 1 220 0 833 1 1 0 833 1 1 0 0025 17 42 1 2 1 2 20 0025 32 ? ? ? ? at at the switching frequency f we choose khz f so c is not needed the power ulator gain at f is gg rfl rfl f f s mhz h mhz h khz khz s zesr f c mod f mc load s load s pmod c c << = = + = 1 5 100 12 21 0 833 1 1 0 833 1 1 17 42 100 0 / ( ). , . mod : () () . .( ) .( ) . . () ? ? 967 967 25 110 0 8 937 33 25 3 11 25 3 11 20 33 then r v gvg v sv k and c v i fl v i fl c r v a mhz h v a mhz h f c out mea fb mod f c out out max s out out max s out c c : . .. : () () . () . () () () () = = = + = + ? k k pf ? 270 figure 6. numerical example to calculate r c and c c values of the typical operating circuit of figure 1 (MAX1953)
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller ______________________________________________________________________________________ 21 1 2 3 4 5 10 9 8 7 6 bst lx dh pgnd gnd fb comp hsd max1954eub max top view dl in 1 2 3 4 5 10 9 8 7 6 bst lx dh pgnd gnd fb comp refin max1957eub max dl in pin configurations (continued)
MAX1953/max1954/max1957 low-cost, high-frequency, current-mode pwm buck controller maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a maxim product. no circu it patent licenses are implied. maxim reserves the right to change the circuitry and specifications without notice at any time. 22 ____________________maxim integrated products, 120 san gabriel drive, sunnyvale, ca 94086 408-737-7600 ? 2002 maxim integrated products printed usa is a registered trademark of maxim integrated products. maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a maxim product. no circu it patent licenses are implied. maxim reserves the right to change the circuitry and specifications without notice at any time. 22 ____________________maxim integrated products, 120 san gabriel drive, sunnyvale, ca 94086 408-737-7600 ? 2002 maxim integrated products printed usa is a registered trademark of maxim integrated products. package information (the package drawing(s) in this data sheet may not reflect the most current specifications. for the latest package outline info rmation, go to www.maxim-ic.com/packages .) 10lumax.eps package outline, 10l umax/usop 1 1 21-0061 i rev. document control no. approval proprietary information title: top view front view 1 0.498 ref 0.0196 ref s 6 side view bottom view 0 0 6 0.037 ref 0.0078 max 0.006 0.043 0.118 0.120 0.199 0.0275 0.118 0.0106 0.120 0.0197 bsc inches 1 10 l1 0.0035 0.007 e c b 0.187 0.0157 0.114 h l e2 dim 0.116 0.114 0.116 0.002 d2 e1 a1 d1 min - a 0.940 ref 0.500 bsc 0.090 0.177 4.75 2.89 0.40 0.200 0.270 5.05 0.70 3.00 millimeters 0.05 2.89 2.95 2.95 - min 3.00 3.05 0.15 3.05 max 1.10 10 0.60.1 0.60.1 ? 0.500.1 h 4x s e d2 d1 b a2 a e2 e1 l l1 c gage plane a2 0.030 0.037 0.75 0.95 a1

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