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description the opa651 is a low power, wideband voltage feed- back operational amplifier. it features a bandwidth at g = +2 of 470mhz as well as a 12-bit settling time of only 16ns. the wide bandwidth and true differential input stage make it suitable for use in a variety of applications. its low distortion gives exceptional per- formance for telecommunications, medical imaging and video applications. the opa651 is compensated for stability in gains of two or more, differentiating it from the unity gain stable opa650. its unusual combination of speed, accuracy and low power make it an outstanding choice for many portable, multi-channel and other high speed applications, where power is at a premium. wideband, low power voltage feedback operational amplifier ? 1994 burr-brown corporation pds-1265e printed in u.s.a. march, 1998 opa651 features l stable in gains: 3 2v/v l low power: 50mw l bandwidth: 470mhz at g = 2 l fast settling time: 16ns to 0.01% l low harmonics: C78db at 5mhz l low input bias current: 4 m a l differential gain/phase error: 0.01%/0.025 l low voltage noise: 4.6nv/ ? hz applications l high resolution video l monitor preamplifier l ccd imaging amplifier l ultrasound signal processing l adc/dac gain amplifier l baseband signal processing current mirror output stage c c inverting input non-inverting input +v s output ? s 1.5pf international airport industrial park ? mailing address: po box 11400, tucson, az 85734 ? street address: 6730 s. tucson bl vd., tucson, az 85706 ? tel: (520) 746-1111 ? twx: 910-952-1111 internet: http://www.burr-brown.com/ ? faxline: (800) 548-6133 (us/canada only) ? cable: bbrcorp ? telex: 066-6491 ? fa x: (520) 889-1510 ? immediate product info: (800) 548-6132 opa651 sbos042
2 opa651 specifications at t a = +25 c, v s = 5v, r l = 100 w , r fb = 402 w, g = +2 , unless otherwise noted. opa651u, n opa651ub, nb parameter conditions min typ max min typ max units frequency response closed-loop bandwidth (2) g = +2 470 * (1) mhz g = +5 70 * mhz g = +10 34 * mhz gain bandwidth product 340 * mhz slew rate g = +2, v o = 2v step 300 * v/ m s over specified temperature 275 * v/ m s rise time 0.2v step 0.8 * ns fall time 0.2v step 0.8 * ns settling time 0.01% g = +2, v o = 2v step 16.2 * ns 0.1% g = +2, v o = 2v step 11.5 * ns 1% g = +2, v o = 2v step 7.2 * ns spurious free dynamic range g = +2, f = 5.0 mhz, v o = 2vp-p, r l = 100 w 67 * dbc r l = 400 w 78 * dbc differential gain g = +2, ntsc, v o = 1.4vp, r l = 150 w 0.01 * % differential phase g = +2, ntsc, v o = 1.4vp, r l = 150 w 0.025 * degrees bandwidth for 0.1db flatness g = +2 43 * mhz input offset voltage input offset voltage 1 5.0 * 2.0 mv average drift 3* m v/ c power supply rejection (+v s )v s = 4.5v to 5.5v 65 85 70 * db (Cv s )526055*db input bias current input bias current v cm = 0v 4 20 * 10 m a over temperature 630 *20 m a input offset current v cm = 0v 0.4 1.5 * * m a over temperature 0.9 3.0 * * m a input noise input voltage noise noise density, f = 100hz 13 * nv/ ? hz f = 10khz 4.6 * nv/ ? hz f = 1mhz 4.6 * nv/ ? hz voltage noise, bw = 10hz to 100mhz 46 * m vrms input bias current noise current noise density, f = 0.1hz to 20khz 1.1 * pa/ ? hz noise figure (nf) r s = 10k w 3.2 * db r s = 50 w 14 * db input voltage range common-mode input range 3.5 * v over specified temperature 3.0 * v common-mode rejection v cm = 0.5v 75 90 80 * db input impedance differential 60 || 1 * k w || pf common-mode 2.6 || 1 * m w || pf open-loop gain open-loop voltage gain v o = 2v, r l = 100 w 42 50 45 * db over specified temperature v o = 2v, r l = 100 w 40 42 db output voltage output over specified temperature no load 2.2 3.0 2.4 * v r l = 250 w 2.2 2.5 2.4 * v r l = 100 w 2.0 2.3 2.1 * v current output, sourcing 75 110 * * ma over specified temperature 65 * ma current output, sinking 65 85 * * ma over specified temperature 35 * ma short circuit current 150 * ma output resistance 0.1mhz, g = +2 0.05 * w power supply specified operating voltage 5*v operating voltage range 4.5 5.5 * * v quiescent current 5.1 7.75 * 6.5 ma over specified temperature 8.75 7.5 ma temperature range specification: u, n, ub, nb C40 +85 * * c thermal resistance, q ja u so-8 125 * c/w n sot23-5 150 * c/w notes: (1) an asterisk (*) specifies the same value as the grade to the left. (2) frequency response can be strongly influenced by pc board parasitics. the opa651 is nominally compensated assuming 2pf parasitic load. the demonstration boards show low parasitic layouts for the different pac kage styles.
3 opa651 1 2 3 5 4 +v s ?nput output ? s +input 1 2 3 4 8 7 6 5 nc +v s output nc nc ?nput +input ? s pin configuration top view so-8 supply ............................................................................................... 5.5v internal power dissipation (1) : .............................. see thermal conditions differential input voltage .................................................................. 1.2v input voltage range ............................................................................ v s storage temperature range: u, ub, n, nb ................ C40 c to +125 c lead temperature (soldering, 10s) .............................................. +300 c (soldering, so-8 3s) ...................................................................... +260 c junction temperature (t j ) ............................................................ +175 c absolute maximum ratings sot23-5 electrostatic discharge sensitivity electrostatic discharge can cause damage ranging from per- formance degradation to complete device failure. burr-brown corporation recommends that all integrated circuits be handled and stored using appropriate esd protection methods. esd damage can range from subtle performance degradation to complete device failure. precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet published speci- fications. package drawing temperature package ordering product package number (1) range marking (2) number (3) opa651u so-8 surface mount 182 C40 c to +85 c opa651u opa651u opa651ub so-8 surface mount 182 C40 c to +85 c opa651ub opa651ub opa651n 5-pin sot23-5 331 C40 c to +85 c a51 OPA651N-250 opa651n-3k opa651nb 5-pin sot23-5 331 C40 c to +85 c a51b opa651nb-250 opa651nb-3k note: (1) for detailed drawing and dimension table, please see end of data sheet, or appendix c of burr-brown ic data book. (2) the b grade of the so-8 package will be marked with a b by pin 8. the b grade of the sot23-5 will be marked with a b near pins 3 and 4. (3) the sot23-5 i s only available on a 7" tape and reel (e.g. ordering 250 pieces of OPA651N-250 will get a single 250 piece tape and reel. ordering 3000 pieces of opa651n-3k will get a single 3000 piece tape and reel). please refer to appendix b of burr-brown ic data book for detailed tape and reel mechanical information. package/ordering information the information provided herein is believed to be reliable; however, burr-brown assumes no responsibility for inaccuracies or o missions. burr-brown assumes no responsibility for the use of this information, and all use of such information shall be entirely at the users own risk. pr ices and specifications are subject to change without notice. no patent rights or licenses to any of the circuits described herein are implied or granted to any third party. burr-brown does not authorize or warrant any burr-brown product for use in life support devices and/or systems.
4 opa651 typical performance curves at t a = +25 c, v s = 5v, r l = 100 w , r fb = 402 w , and g = +2, unless otherwise noted. output current vs temperature 110 100 90 80 ?0 ?5 0 25 50 75 100 output current (?a) temperature (?) i o + i o input voltage noise density vs frequency 100 1k 10k 100k 1m input voltage noise (nv/ ? hz) frequency (hz) 100 10 1 supply current vs temperature 8 7 6 5 4 3 2 1 0 ?5 ?0 ?5 0 25 50 75 100 supply current (?a) ambient temperature (?) input bias current and offset voltage vs temperature 7 6 5 4 2 1 0 ? ?0 ?5 0 25 50 75 100 input bias current (ma) offset voltage (mv) temperature (?) v os i b common-mode rejection ratio vs common-mode input voltage 100 90 80 70 60 50 40 30 ? ? 3 101234 common-mode rejection (db) common-mode voltage (v) a ol , psr+, psr? and cmrr vs temperature 100 95 90 85 80 75 70 65 60 55 50 45 40 ?5 ?0 ?5 0 25 50 75 100 a ol , psr+, psr? and cmrr (db) ambient temperature (?) a ol psr+ psr cmrr
5 opa651 typical performance curves (cont) at t a = +25 c, v s = 5v, r l = 100 w , r fb = 402 w , and g = +2, unless otherwise noted. small signal bandwidth g = +10 closed-loop 100m frequency (hz) 1m 0 10m 1g 23 20 17 14 11 8 5 gain (db) bandwidth = 34mhz large signal transient response (g = +2) time (5ns/div) 2.0 1.6 1.2 0.8 0.4 0 ?.4 ?.8 ?.2 ?.6 ?.0 output voltage (v) small signal transient response (g = +2) time (5ns/div) 160 120 80 40 0 ?0 ?0 ?20 ?60 output voltage (mv) recommended isolation resistance vs capacitive load 40 30 20 10 0 0 20 40 60 80 100 isolation resistance, r iso ( w ) capacitive load, c l (pf) r iso opa651 c l 1k w 402 w 402 w small signal bandwidth g = +2 closed-loop 100m frequency (hz) 1m 0 10m 1g 12 9 6 3 0 ? gain (db) so-8 bandwidth = 472mhz small signal bandwidth g = +5 closed-loop 100m frequency (hz) 1m 0 10m 1g 20 17 14 11 8 5 2 gain (db) so-8 bandwidth = 66mhz
6 opa651 typical performance curves (cont) at t a = +25 c, v s = 5v, r l = 100 w , r fb = 402 w , and g = +2, unless otherwise noted. harmonic distortion vs gain (f = 5mhz, v o = 2v p? ) non-inverting gain (v/v) 12345678910 ?0 ?0 ?0 ?0 ?0 harmonic distortion (dbc) 3fo 2fo 10mhz harmonic distortion vs output swing (g = +2) output swing (v p? ) 01 234 ?0 ?5 ?0 ?5 ?0 ?5 ?0 ?5 ?0 harmonic distortion (dbc) 3fo 2fo 5mhz harmonic distortion vs output swing (g = +2) output swing (v p? ) 01 234 ?5 ?0 ?5 ?0 ?5 ?0 ?5 ?0 ?5 harmonic distortion (dbc) 3fo 2fo harmonic distortion vs temperature (f = 5mhz, v o = 2v p-p , g = +2) ?5 ?0 ?5 ?0 ?5 ?5 ?0 ?5 0 25 50 75 100 harmonic distortion (dbc) ambient temperature (?) 3fo 2fo harmonic distortion vs frequency (g = +2, v o = 2v p?, r l = 100 w ) 10m frequency (hz) 100k 1m 100m ?0 ?5 ?0 ?5 ?0 ?5 ?0 ?5 ?0 ?5 ?0 harmonic distortion (dbc) 3fo 2fo open-loop gain and phase vs frequency 60 40 20 0 ?0 ?0 0 ?5 ?0 ?35 ?80 ?25 1k 10k 100k 1m 10m 100m 1g gain (db) phase (? frequency (hz) gain phase
7 opa651 discussion of performance the opa651 is a low power, wideband voltage feedback operational amplifier, internally compensated to provide gain of +2 stability. the opa651s voltage feedback architecture features true differential and fully symmetrical inputs. this minimizes offset errors, making the opa651 well suited for implementing filter and instrumentation designs. the opa651s ac performance is optimized to provide a gain bandwidth product of 340mhz and a fast 0.1% settling time of 11.5ns, which is an important consideration in high speed data conversion applications. along with its excellent settling characteristics, the low dc input offset of 1mv and drift of 3 m v/ c support high accuracy requirements. in applica- tions requiring a higher slew rate and wider bandwidth, such as video and high bit rate digital communications, consider the current feedback opa658. circuit layout and basic operation achieving optimum performance with a high frequency am- plifier like the opa651 requires careful attention to layout parasitics and selection of external components. recommen- dations for pc board layout and component selection include: a) minimize parasitic capacitance to any ac ground for all of the signal i/o pins. parasitic capacitance on the output and inverting input pins can cause instability; on the non- inverting input it can react with the source impedance to cause unintentional bandlimiting. to reduce unwanted ca- pacitance, a window around the signal i/o pins should be opened in all of the ground and power planes. otherwise, ground and power planes should be unbroken elsewhere on the board. b) minimize the distance (< 0.25") from the two power pins to high frequency 0.1 m f decoupling capacitors. at the pins, the ground and power plane layout should not be in close proximity to the signal i/o pins. avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. larger (2.2 m f to 6.8 m f) decoupling capacitors, effective at lower frequencies, should also be used. these may be placed somewhat farther from the device and may be shared among several devices in the same area of the pc board. c) careful selection and placement of external compo- nents will preserve the high frequency performance of the opa651 . resistors should be a very low reactance type. surface mount resistors work best and allow a tighter overall layout. metal film or carbon composition axially-leaded resistors can also provide good high frequency performance. again, keep their leads as short as possible. never use wirewound type resistors in a high frequency application. since the output pin and the inverting input pin are most sensitive to parasitic capacitance, always position the feed- back and series output resistor, if any, as close as possible to the package pins. other network components, such as non- inverting input termination resistors, should also be placed close to the package. even with a low parasitic capacitance shunting external resistors, excessively high resistor values can create signifi- cant time constants and degrade performance. this is par- ticularly true for the opa651 with its internal 1.5pf feed- back capacitance. good metal film or surface mount resis- tors have approximately 0.2pf in shunt with the resistor. for resistor values > 1.5k w , this adds a pole and/or zero below 500mhz that can affect circuit operation. keep resistor values as low as possible consistent with output loading considerations. the 402 w feedback used for the typical performance plots is a good starting point for design. d) connections to other wideband devices on the board may be made with short direct traces or through on-board transmission lines. for short connections, consider the trace and the input to the next device as a lumped capacitive load. relatively wide traces (50 to 100 mils) should be used, preferably with ground and power planes opened up around them. estimate the total capacitive load and set r iso from the plot of recommended r iso vs capacitive load. low parasitic loads may not need an r iso since the opa651 is nominally compensated to operate with a 2pf parasitic load. if a long trace is required and the 6db signal loss intrinsic to doubly terminated transmission lines is acceptable, imple- ment a matched impedance transmission line using microstrip or stripline techniques (consult an ecl design handbook for microstrip and stripline layout techniques). a 50 w environ- ment is not necessary on board, and in fact a higher imped- ance environment will improve distortion as shown in the distortion vs load plot. with a characteristic impedance defined based on board material and desired trace dimen- sions, a matching series resistor into the trace from the output of the amplifier is used as well as a terminating shunt resistor at the input of the destination device. remember also that the terminating impedance will be the parallel combination of the shunt resistor and the input impedance of the destination device; the total effective impedance should match the trace impedance. multiple destination devices are best handled as separate transmission lines, each with their own series and shunt terminations. if the 6db attenuation loss of a doubly terminated line is unacceptable, a long trace can be series-terminated at the source end only. this will help isolate the line capacitance from the op amp output, but will not preserve signal integrity as well as a doubly terminated line. if the shunt impedance at the destination end is finite, there will be some signal attenuation due to the voltage divider formed by the series and shunt impedances. the opa651 is nominally specified for operation using 5v power supplies. a 10% tolerance on the supplies, or an ecl C5.2v for the negative supply, is within the maximum speci- fied total supply voltage of 11v. higher supply voltages can break down internal junctions possibly leading to catastrophic failure. single supply operation is possible as long as com- mon mode voltage constraints are observed. the common
8 opa651 mode input and output voltage specifications can be inter- preted as a required headroom to the supply voltage. observ- ing this input and output headroom requirement will allow non-standard or single supply operation. figure 1 shows one approach to single-supply operation. esd protection esd damage has been well recognized for mosfet de- vices, but any semiconductor device is vulnerable to this potentially damaging source. this is particularly true for very high speed, fine geometry processes. esd damage can cause subtle changes in amplifier input characteristics without necessarily destroying the device. in precision operational amplifiers, this may cause a noticeable degradation of offset voltage and drift. therefore, esd handling precautions are strongly recommended when han- dling the opa651. output drive capability the opa651 has been optimized to drive 75 w and 100 w resistive loads. the device can drive a 2vp-p into a 75 w load. this high-output drive capability makes the opa651 an ideal choice for a wide range of rf, if, and video applications. in many cases, additional buffer amplifiers are unneeded. many demanding high-speed applications such as driving a/d converters require op amps with low wideband output impedance. for example, low output impedance is essential when driving the signal-dependent capacitances at the inputs of flash a/d converters. as shown in figure 3, the opa651 maintains very low-closed loop output impedance over fre- quency. closed-loop output impedance increases with fre- quency since loop gain is decreasing. 402 w opa651 v ac 402 w r l +v s +v s v s 2 r out v s 2 v out = + 2? ac thermal considerations the opa651 will not require heatsinking under most oper- ating conditions. maximum desired junction temperature will limit the maximum allowed internal power dissipation as described below. in no case should the maximum junction temperature be allowed to exceed +175 c. figure 3. small-signal output impedance vs frequency. small-signal output impedance vs frequency frequency (hz) 1k 100 10 1 0.1 0.01 10k 100k 1m 100m 10m output impedance ( w ) g = +2 offset voltage adjustment if additional offset adjustment is needed, the circuit in figure 2 can be used without degrading offset drift with temperature. avoid external adjustment whenever possible since extraneous noise, such as power supply noise, can be inadvertently coupled into the amplifiers inverting input terminal. remember that additional offset errors can be created by the amplifiers input bias currents. whenever possible, match the impedance seen by both inputs as is shown with r 3 . this will reduce input bias current errors to the amplifiers offset current. r 2 opa651 (1) r 3 = r 1 || r 2 r 1 r trim +v s ? s 20k w v in or ground output trim range +v s to ? s @ r trim 47k w r 2 r 2 r trim 0.1? note: (1) r 3 is optional and can be used to cancel offset errors due to input bias currents. figure 2. offset voltage trim. figure 1. single supply operation.
9 opa651 operating junction temperature (t j ) is given by t a + p d q ja . the total internal power dissipation (p d ) is a com- bination of the total quiescent power (p dq ) and the power dissipated in of the output stage (p dl ) to deliver load power. quiescent power is simply the specified no-load supply current times the total supply voltage across the part. p dl will depend on the required output signal and load but would, for a grounded resistive load, be at a maximum when the output is a fixed dc voltage equal to 1/2 of either supply voltage (assuming equal bipolar supplies). under this condition, p dl = v s 2 /(4?r l ) where r l includes feed- back network loading. note that it is the power dissipated in the output stage and not in the load that determines internal power dissipation. as an example, compute the maximum t j for an opa651n at a v = +2, r l = 100 w , r fb = 402 w , v s = 5v, with the output at |v s /2|, and the specified maximum t a = +85 c. p d = 10v?8.75ma + (5 2 )/ (4?(100 w ||804 w )) = 158mw. maximum t j = +85 c + 0.158w?150 c/w = 109 c. driving capacitive loads the opa651s output stage has been optimized to drive low resistive loads. capacitive loads, however, will decrease the amplifiers phase margin which may cause high frequency peaking or oscillations. capacitive loads greater than 10pf should be isolated by connecting a small resistance, usually 15 w to 30 w , in series with the output as shown in figure 4. this is particularly important when driving high capacitance loads such as flash a/d converters. increasing the gain from +2 will improve the capacitive load drive due to increased phase margin. in general, capacitive loads should be minimized for opti- mum high frequency performance. coax lines can be driven if the cable is properly terminated. the capacitance of coax cable (29pf/foot for rg-58) will not load the amplifier when the coaxial cable or transmission line is terminated in its characteristic impedance. that, from a stability standpoint, an inverting gain of C1v/v is equivalent to a noise gain of 2.) frequency response for other gains are shown in the typical performance curves. the high frequency response of the opa651 in a good layout is very flat with frequency. however, some circuit configurations such as those where large feedback resis- tances are used, can produce high-frequency gain peaking. this peaking can be minimized by connecting a small capacitor in parallel with the feedback resistor. this capaci- tor compensates for the closed-loop, high-frequency, trans- fer function zero that results from the time constant formed by the input capacitance of the amplifier (typically 2pf after pc board mounting), and the input and feedback resistors. the selected compensation capacitor may be a trimmer, a fixed capacitor, or a planned pc board capacitance. the capacitance value is strongly dependent on circuit layout and closed-loop gain. using small resistor values will preserve the phase margin and avoid peaking by keeping the break frequency of this zero sufficiently high. when high closed- loop gains are required, a three-resistor attenuator (tee- network) is recommended to avoid using large value resis- tors with large time constants. the opa651 includes an internal 1.5pf feedback capacitor to achieve best gain of +2 flatness (r f = 402 w ). pulse settling time high speed amplifiers like the opa651 are capable of extremely fast settling time with a pulse input. excellent frequency response flatness and phase linearity are required to get the best settling times. as shown in the specifications table, settling time for a 1v step at a gain of +2 for the opa651 is extremely fast. the specification is defined as the time required, after the input transition, for the output to settle within a specified error band around its final value. for a 2v step, 1% settling corresponds to an error band of 20mv, 0.1% to an error band of 2mv, and 0.01% to an error band of 0.2mv. for the best settling times, particu- larly into an adc capacitive load, little or no peaking in the frequency response can be allowed. using the recommended r iso for capacitive loads will limit this peaking and reduce the settling times. fast, extremely fine scale settling (0.01%) requires close attention to ground return currents in the supply decoupling capacitors. for highest performance, con- sider the opa642 which isolates the output stage decoupling from the rest of the amplifier. differential gain and phase differential gain (dg) and differential phase (dp) are among the more important specifications for video applica- tions. the percentage change in closed-loop gain over a specified change in output voltage level is defined as dg. dp is defined as the change in degrees of the closed-loop phase over the same output voltage change. dg and dp are both specified at the ntsc sub-carrier frequency of 3.58mhz. all measurements were performed using an hp 9480. figure 4. driving capacitive loads. opa651 c l r l r iso (r iso typically 15 w to 30 w ) 402 w 402 w frequency response compensation the opa651 is internally compensated and is stable at a gain of 2 with a phase margin of approximately 60 . (note
10 opa651 distortion the opa651s harmonic distortion characteristics into a 100 w load are shown versus frequency and power output in the typical performance curves. distortion can be signifi- cantly improved by increasing the load resistance as illus- trated in figure 5. remember to include the contribution of the feedback network when calculating the effective load resistance seen by the amplifier. 10 100 1k ?0 ?5 ?0 ?5 ?0 ?5 ?0 harmonic distortion (dbc) 3fo 2fo g = +2 load resistance ( w ) figure 5. 5mhz harmonic distortion vs load resistance. noise figure the opa651 voltage noise spectral density is specified in the typical performance curves. for rf applications, how- ever, noise figure (nf) is often the preferred noise specifi- cation since it allows system noise performance to be more easily calculated. the opa651s noise figure vs source resistance is shown in figure 6. noise figure vs source resistance source resistance ( w ) 30 25 20 15 10 5 0 10 100 1k 100k 10k noise figure (db) nf = 10 log 1 + e n 2 + (i n r s ) 2 4ktr s figure 6. noise figure vs source resistance. spice models and evaluation board computer simulation of circuit performance using spice is often useful when analyzing the performance of analog circuits and systems. this is particularly true for video and rf amplifier circuits where parasitic capacitance and induc- tance can have a major effect on circuit performance. spice models are available on a disk from the burr-brown appli- cations department. demonstration boards are available for each opa651 pack- age style. these boards implement a very low parasitic layout that will produce the excellent frequency and pulse responses shown in the typical performance curves. for each package style, the recommended demonstration board is: contact your local burr-brown sales office or distributor to order demonstration boards. board literature part request product package number number opa651p 8-pin dip dem-opa65xp opa651u 8-pin so-8 dem-opa65xu opa651n 6-pin sot23-6 dem-opa6xxn mkt-348 opa651 v out 402 w 402 w video input 75 w 75 w 75 w transmission line 75 w typical application figure 7. low distortion video amplifier.
11 opa651 figure 8. layout detail for dem-opa65xu demonstration board. note: values for r 1 , r 2 , r 3 , r 5 , r 6 , r 7 and r 8 are chosen according to desired gain. r 6 r 1 opa651 out j 1 1 2 gnd ?v p2 r 7 r 5 +in r 3 r 4 r 2 ?n j 1 j 2 2 7 4 6 3 c 2 0.1? c 4 2.2? 402 w c 1 2.2? c 3 0.1? 1 2 gnd +5v p1 r 8 + +
important notice texas instruments and its subsidiaries (ti) reserve the right to make changes to their products or to discontinue any product or service without notice, and advise customers to obtain the latest version of relevant information to verify, before placing orders, that information being relied on is current and complete. all products are sold subject to the terms and conditions of sale supplied at the time of order acknowledgment, including those pertaining to warranty, patent infringement, and limitation of liability. ti warrants performance of its semiconductor products to the specifications applicable at the time of sale in accordance with ti's standard warranty. testing and other quality control techniques are utilized to the extent ti deems necessary to support this warranty. specific testing of all parameters of each device is not necessarily performed, except those mandated by government requirements. customers are responsible for their applications using ti components. in order to minimize risks associated with the customer's applications, adequate design and operating safeguards must be provided by the customer to minimize inherent or procedural hazards. ti assumes no liability for applications assistance or customer product design. ti does not warrant or represent that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other intellectual property right of ti covering or relating to any combination, machine, or process in which such semiconductor products or services might be or are used. ti's publication of information regarding any third party's products or services does not constitute ti's approval, warranty or endorsement thereof. copyright ? 2000, texas instruments incorporated

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