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LT3757 Boost, Flyback, SEPIC and Inverting Controller FeaTures
Wide Input Voltage Range: 2.9V to 40V n Positive or Negative Output Voltage Programming with a Single Feedback Pin n CurrentModeControlProvidesExcellentTransient Response n ProgrammableOperatingFrequency(100kHzto 1MHz)withOneExternalResistor n SynchronizabletoanExternalClock n LowShutdownCurrent<1A n Internal7.2VLowDropoutVoltageRegulator n ProgrammableInputUndervoltageLockoutwith Hysteresis n ProgrammableSoft-Start n Small10-LeadDFN(3mmx3mm)andThermally Enhanced10-PinMSOPPackages
n
DescripTion
TheLT(R)3757isawideinputrange,currentmode,DC/DC controllerwhichiscapableofgeneratingeitherpositiveor negativeoutputvoltages.Itcanbeconfiguredaseithera boost,flyback,SEPICorinvertingconverter.TheLT3757 drivesalowsideexternalN-channelpowerMOSFETfrom an internal regulated 7.2V supply. The fixed frequency, current-modearchitectureresultsinstableoperationover awiderangeofsupplyandoutputvoltages. The operating frequency of LT3757 can be set with an externalresistorovera100kHzto1MHzrange,andcan besynchronizedtoanexternalclockusingtheSYNCpin. Alowminimumoperatingsupplyvoltageof2.9V,anda lowshutdownquiescentcurrentoflessthan1A,make theLT3757ideallysuitedforbattery-operatedsystems. The LT3757 features soft-start and frequency foldback functions to limit inductor current during start-up and outputshort-circuit.
L,LT,LTC,LTM,LinearTechnology,theLinearlogoandBurstModeareregisteredtrademarks andNoRSENSEandThinSOTaretrademarksofLinearTechnologyCorporation.Allother trademarksarethepropertyoftheirrespectiveowners.
applicaTions
AutomotiveandIndustrialBoost,Flyback,SEPICand InvertingConverters n TelecomPowerSupplies n PortableElectronicEquipment
n
Typical applicaTion
High Efficiency Boost Converter
VIN 8V TO 16V 10F 25V X5R 100 200k VIN SHDN/UVLO 43.2k SYNC RT SS 41.2k 300kHz 0.1F VC 22k 6.8nF 10H VOUT 24V 2A 226k 90 80 EFFICIENCY (%) 70 60 50 10F 25V X5R 40 30 0.001 0.1 1 0.01 OUTPUT CURRENT (A) 10
3757 TA01b 3757 TA01a
Efficiency
VIN = 8V VIN = 16V
LT3757
GATE SENSE
FBX GND INTVCC 4.7F 10V X5R
+
16.2k 0.01
47F 35V 2
3757fb
LT3757 absoluTe MaxiMuM raTings
(Note 1)
VIN,SHDN/UVLO(Note6).........................................40V INTVCC....................................................VIN+0.3V,20V GATE........................................................ INTVCC+0.3V SYNC..........................................................................8V . VC,SS ........................................................................3V RT............................................................................1.5V SENSE ...................................................................0.3V . FBX................................................................. -6Vto6V
OperatingTemperatureRange(Notes2,8) LT3757E............................................ -40Cto125C . LT3757I............................................. -40Cto125C . LT3757H............................................ -40Cto150C LT3757MP......................................... -55Cto125C StorageTemperatureRange DFN................................................... -65Cto125C . MSOP................................................ -65Cto150C LeadTemperature(Soldering,10sec) MSOP............................................................... 300C
pin conFiguraTion
TOP VIEW VC FBX SS RT SYNC 1 2 3 4 5 11 10 VIN 9 SHDN/UVLO 8 INTVCC 7 GATE 6 SENSE TOP VIEW VC FBX SS RT SYNC 1 2 3 4 5 10 9 8 7 6 VIN SHDN/UVLO INTVCC GATE SENSE
11
DD PACKAGE 10-LEAD (3mm 3mm) PLASTIC DFN TJMAX=125C,JA=43C/W EXPOSEDPAD(PIN11)ISGND,MUSTBESOLDEREDTOPCB
MSE PACKAGE 10-LEAD PLASTIC MSOP TJMAX=150C,JA=40C/W EXPOSEDPAD(PIN11)ISGND,MUSTBESOLDEREDTOPCB
orDer inForMaTion
LEAD FREE FINISH LT3757EDD#PBF LT3757IDD#PBF LT3757EMSE#PBF LT3757IMSE#PBF LT3757HMSE#PBF LT3757MPMSE#PBF TAPE AND REEL LT3757EDD#TRPBF LT3757IDD#TRPBF LT3757EMSE#TRPBF LT3757IMSE#TRPBF LT3757HMSE#TRPBF LT3757MPMSE#TRPBF PART MARKING* LDYW LDYW LTDYX LTDYX LTDYX LTDYX PACKAGE DESCRIPTION 10-Lead(3mmx3mm)PlasticDFN 10-Lead(3mmx3mm)PlasticDFN 10-Lead(3mmx3mm)PlasticMSOP 10-Lead(3mmx3mm)PlasticMSOP 10-Lead(3mmx3mm)PlasticMSOP 10-Lead(3mmx3mm)PlasticMSOP TEMPERATURE RANGE -40Cto125C -40Cto125C -40Cto125C -40Cto125C -40Cto150C -55Cto125C
ConsultLTCMarketingforpartsspecifiedwithwideroperatingtemperatureranges.*Thetemperaturegradeisidentifiedbyalabelontheshippingcontainer. Formoreinformationonleadfreepartmarking,goto:http://www.linear.com/leadfree/ Formoreinformationontapeandreelspecifications,goto:http://www.linear.com/tapeandreel/
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LT3757 elecTrical characTerisTics
PARAMETER VINOperatingRange VINShutdownIQ VINOperatingIQ VINOperatingIQwithInternalLDODisabled SENSECurrentLimitThreshold SENSEInputBiasCurrent Error Amplifier FBXRegulationVoltage(VFBX(REG)) FBXOvervoltageLockout FBXPinInputCurrent Transconductancegm(IVC /VFBX) VCOutputImpedance VFBXLineRegulation[VFBX /(VIN*VFBX(REG))] VCCurrentModeGain(VVC /VSENSE) VCSourceCurrent VCSinkCurrent Oscillator SwitchingFrequency RT=41.2ktoGND,VFBX=1.6V RT=140ktoGND,VFBX=1.6V RT=10.5ktoGND,VFBX=1.6V VFBX=1.6V 270 300 100 1000 1.2 220 220 0.4 1.5 SS=0V,CurrentOutofPin
l
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 24V, SHDN/UVLO = 24V, SENSE = 0V, unless otherwise noted.
CONDITIONS SHDN/UVLO=0V SHDN/UVLO=1.15V VC=0.3V,RT=41.2k VC=0.3V,RT=41.2k,INTVCC=7.5V
l
MIN 2.9
TYP 0.1 1.6 280
MAX 40 1 6 2.2 400 120
UNITS V A A mA A mV A
100
110 -65
CurrentOutofPin VFBX>0V(Note3) VFBX<0V(Note3) VFBX>0V(Note4) VFBX<0V(Note4) VFBX=1.6V(Note3) VFBX=-0.8V(Note3) (Note3) (Note3) VFBX>0V,2.9Vl l
1.569 -0.816 6 7 -10
1.6 -0.80 8 11 70 230 5 0.002 0.0025 5.5 -15 12 11
1.631 -0.784 10 14 100 10
V V % % nA nA S M
0.056 0.05
%/V %/V V/V A A A
330
kHz kHz kHz V ns ns V V A
RTVoltage MinimumOff-Time MinimumOn-Time SYNCInputLow SYNCInputHigh SSPull-UpCurrent Low Dropout Regulator INTVCCRegulationVoltage INTVCCUndervoltageLockoutThreshold INTVCCOvervoltageLockoutThreshold INTVCCCurrentLimit INTVCCLoadRegulation(VINTVCC / VINTVCC) INTVCCLineRegulationVINTVCC /(VINTVCC*VIN) DropoutVoltage(VIN-VINTVCC)
-10 7 2.6 16 7.2 2.7 0.1 17.5 40 95 -0.5 0.008 400 0.03 55 7.4 2.8
V V V V mA mA % %/V mV
FallingINTVCC UVLOHysteresis VIN=40V VIN=15V 030 -0.9
3757fb
LT3757 elecTrical characTerisTics
PARAMETER INTVCCCurrentinShutdown INTVCCVoltagetoBypassInternalLDO Logic Inputs SHDN/UVLOThresholdVoltageFalling SHDN/UVLOInputLowVoltage SHDN/UVLOPinBiasCurrentLow SHDN/UVLOPinBiasCurrentHigh Gate Driver t rGateDriverOutputRiseTime t fGateDriverOutputFallTime GateVOL GateVOH Note 1:StressesbeyondthoselistedunderAbsoluteMaximumRatings maycausepermanentdamagetothedevice.ExposuretoanyAbsolute MaximumRatingconditionforextendedperiodsmayaffectdevice reliabilityandlifetime. Note 2: TheLT3757Eisguaranteedtomeetperformancespecifications fromthe0Cto125Cjunctiontemperature.Specificationsoverthe-40C to125Coperatingjunctiontemperaturerangeareassuredbydesign, characterizationandcorrelationwithstatisticalprocesscontrols.The LT3757Iisguaranteedoverthefull-40Cto125Coperatingjunction temperaturerange.TheLT3757Hisguaranteedoverthefull-40Cto150C operatingjunctiontemperaturerange.Highjunctiontemperaturesdegrade operatinglifetimes.Operatinglifetimeisderatedatjunctiontemperatures greaterthan125C.TheLT3757MPis100%testedandguaranteedoverthe full-55Cto125Coperatingjunctiontemperaturerange. INTVCC -0.05 CL=3300pF(Note5),INTVCC=7.5V CL=3300pF(Note5),INTVCC=7.5V 22 20 0.05 ns ns V V VIN=INTVCC=8V I(VIN)DropsBelow1A SHDN/UVLO=1.15V SHDN/UVLO=1.30V 1.7 2 10
l
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 24V, SHDN/UVLO = 24V, SENSE = 0V, unless otherwise noted.
CONDITIONS SHDN/UVLO=0V,INTVCC=8V MIN TYP 16 7.5 1.17 1.22 1.27 0.4 2.5 100 MAX UNITS A V V V A nA
Note 3:TheLT3757istestedinafeedbackloopwhichservosVFBXtothe referencevoltages(1.6Vand-0.8V)withtheVCpinforcedto1.3V. Note 4:FBXovervoltagelockoutismeasuredatVFBX(OVERVOLTAGE)relative toregulatedVFBX(REG). Note 5:Riseandfalltimesaremeasuredat10%and90%levels. Note 6:ForVINbelow6V,theSHDN/UVLOpinmustnotexceedVIN. Note 7:SHDN/UVLO=1.33VwhenVIN=2.9V. Note 8:TheLT3757includesovertemperatureprotectionthatisintended toprotectthedeviceduringmomentaryoverloadconditions.Junction temperaturewillexceedthemaximumoperatingjunctiontemperature whenovertemperatureprotectionisactive.Continuousoperationabove thespecifiedmaximumoperatingjunctiontemperaturemayimpairdevice reliability.
Typical perForMance characTerisTics
Positive Feedback Voltage vs Temperature, VIN
1605 REGULATED FEEDBACK VOLTAGE (mV) VIN = 40V 1600 VIN = 24V REGULATED FEEDBACK VOLTAGE (mV) -788 -790 -792 -794 -796 -798 -800 -802 -804 -75 -50 -25 VIN = 24V VIN = 8V VIN = INTVCC = 2.9V SHDN/UVLO = 1.33V
TA = 25C, unless otherwise noted. Quiescent Current vs Temperature, VIN
1.8
Negative Feedback Voltage vs Temperature, VIN
QUIESCENT CURRENT (mA)
1.7
VIN = 40V
1595 VIN = 8V 1590 VIN = INTVCC = 2.9V SHDN/UVLO = 1.33V
VIN = 24V
1.6
VIN = 40V
1585
1.5 VIN = INTVCC = 2.9V
1580 -75 -50 -25
0 25 50 75 100 125 150 TEMPERATURE (C)
3757 G01
0 25 50 75 100 125 150 TEMPERATURE (C)
3757 G02
1.4 -75 -50 -25
0 25 50 75 100 125 150 TEMPERATURE (C)
3757 G03
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LT3757 Typical perForMance characTerisTics
Dynamic Quiescent Current vs Switching Frequency
35 30 25 RT (k ) IQ(mA) 20 15 10 5 0 0 100 200 300 400 500 600 700 800 900 1000 SWITCHING FREQUENCY (KHz)
3757 G04
TA = 25C, unless otherwise noted. Normalized Switching Frequency vs FBX
120 NORMALIZED FREQUENCY (%) 100 80 60 40 20 0 -0.8
RT vs Switching Frequency
1000
CL = 3300pF
100
10
0 100 200 300 400 500 600 700 800 900 1000 SWITCHING FREQUENCY (KHz)
3757 G05
-0.4
0 0.4 0.8 FBX VOLTAGE (V)
1.2
1.6
3757 G06
Switching Frequency vs Temperature
330 320 SENSE THRESHOLD (mV) 310 300 290 280 270 -75 -50 -25 RT = 41.2K 120
SENSE Current Limit Threshold vs Temperature
115
SENSE Current Limit Threshold vs Duty Cycle
SWITCHING FREQUENCY (kHz)
110
SENSE THRESHOLD (mV)
115
110
105
105
100
0 25 50 75 100 125 150 TEMPERATURE (C)
3757 G07
100 -75 -50 -25
0 25 50 75 100 125 150 TEMPERATURE (C)
3757 G08
95
0
20
40 60 DUTY CYCLE (%)
80
100
3757 G09
SHDN/UVLO Threshold vs Temperature
1.28 40
SHDN/UVLO Current vs Voltage
2.4
SHDN/UVLO Hysteresis Current vs Temperature
SHDN/UVLO CURRENT (A)
SHDN/UVLO VOLTAGE (V)
1.26 SHDN/UVLO RISING
30 ISHDN/ UVLO (A) 0 10 20 30 SHDN/UVLO VOLTAGE (V) 40
3757 G11
2.2
1.24
20
2.0
1.22 SHDN/UVLO FALLING 1.20
10
1.8
1.18 -75 -50 -25
0 25 50 75 100 125 150 TEMPERATURE (C)
3757 G10
0
1.6 -75 -50 -25
0 25 50 75 100 125 150 TEMPERATURE (C)
3757 G12
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LT3757 Typical perForMance characTerisTics
7.4
TA = 25C, unless otherwise noted.
INTVCC vs Temperature
90 80 INTVCC CURRENT (mA)
INTVCC Minimum Output Current vs VIN
TJ = 150C 7.3
INTVCC Load Regulation
VIN = 8V
60 50 40 30 20 10 INTVCC = 6V INTVCC = 4.5V
INTVCC VOLTAGE (V) 30 35 40
7.3 INTVCC (V)
70
7.2
7.1 7
7.2
7.1
6.9
7.0 -75 -50 -25
0 25 50 75 100 125 150 TEMPERATURE (C)
3757 G13
0
0
5
10
15
20 25 VIN (V)
6.8
0
10
3757 G14
40 50 30 20 INTVCC LOAD (mA)
60
70
3757 G15
INTVCC Line Regulation
7.30
INTVCC Dropout Voltage vs Current, Temperature
700 600 DROPOUT VOLTAGE (mV) VIN = 6V 125C 150C 90 80 75C TIME (ns) 25C 0C -55C 70 60 50 40 30 20 100 10 0 5 10 INTVCC LOAD (mA) 15 20
3757 G17
Gate Drive Rise and Fall Time vs CL
INTVCC = 7.2V
INTVCC VOLTAGE (V)
7.25
500 400 300 200
RISE TIME
FALL TIME
7.20
7.15
7.10
0
5
10
15
20 25 VIN (V)
30
35
40
0
0
0
5
10
15 CL (nF)
20
25
30
3757 G18
3757 G16
Gate Drive Rise and Fall Time vs INTVCC
30 25 20 TIME (ns) FALL TIME 15 10 5 0 IL1A + IL1B 5A/DIV CL = 3300pF RISE TIME VOUT 5V/DIV
Typical Start-Up Waveforms
VIN = 12V VOUT 10V/DIV VSW 20V/DIV
FBX Frequency Foldback Waveforms During Overcurrent
VIN = 12V
IL1A + IL1B 5A/DIV
3757 G20 3757 G21
2ms/DIV PAGE 31 CIRCUIT 3 6 9 INTVCC (V)
3757 G19
50s/DIV PAGE 31 CIRCUIT
12
15
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LT3757 pin FuncTions
VC (Pin 1):ErrorAmplifierCompensationPin.Usedto stabilizethevoltageloopwithanexternalRCnetwork. FBX (Pin 2):PositiveandNegativeFeedbackPin.Receives the feedback voltage from the external resistor divider acrosstheoutput.Alsomodulatesthefrequencyduring start-upandfaultconditionswhenFBXisclosetoGND. SS (Pin 3):Soft-StartPin.Thispinmodulatescompensationpinvoltage(VC)clamp.Thesoft-startintervalisset withanexternalcapacitor.Thepinhasa10A(typical) pull-upcurrentsourcetoaninternal2.5Vrail.ThesoftstartpinisresettoGNDbyanundervoltagecondition atSHDN/UVLO,anINTVCCundervoltageorovervoltage conditionoraninternalthermallockout. RT (Pin 4):SwitchingFrequencyAdjustmentPin.Setthe frequencyusingaresistortoGND.Donotleavethispin open. SYNC (Pin 5):FrequencySynchronizationPin.Usedto synchronizetheswitchingfrequencytoanoutsideclock. Ifthisfeatureisused,anRTresistorshouldbechosento programaswitchingfrequency20%slowerthantheSYNC pulsefrequency.TietheSYNCpintoGNDifthisfeatureis notused.SYNCisignoredwhenFBXisclosetoGND. SENSE (Pin 6):TheCurrentSenseInputfortheControl Loop.Kelvinconnectthispintothepositiveterminalof the switch current sense resistor in the source of the N-channelMOSFET.Thenegativeterminalofthecurrent senseresistorshouldbeconnectedtoGNDplaneclose totheIC. GATE (Pin 7): N-Channel MOSFET Gate Driver Output. SwitchesbetweenINTVCCandGND.DriventoGNDwhen ICisshutdown,duringthermallockoutorwhenINTVCCis aboveorbelowtheOVorUVthresholds,respectively. INTVCC (Pin 8):RegulatedSupplyforInternalLoadsand GateDriver.SuppliedfromVINandregulatedto7.2V(typical).INTVCCmustbebypassedwithaminimumof4.7F capacitorplacedclosetopin.INTVCCcanbeconnected directlytoVIN,ifVINislessthan17.5V.INTVCCcanalso beconnectedtoapowersupplywhosevoltageishigher than7.5V,andlowerthanVIN,providedthatsupplydoes notexceed17.5V. SHDN/UVLO (Pin 9):ShutdownandUndervoltageDetect Pin.Anaccurate1.22V(nominal)fallingthresholdwith externallyprogrammablehysteresisdetectswhenpower isokaytoenableswitching.Risinghysteresisisgenerated bytheexternalresistordividerandanaccurateinternal 2Apull-downcurrent.Anundervoltageconditionresets sort-start.Tieto0.4V,orless,todisablethedeviceand reduceVINquiescentcurrentbelow1A. VIN (Pin 10): InputSupplyPin.Mustbelocallybypassed with a 0.22F or larger, capacitor placed close to the , pin. Exposed Pad (Pin 11):Ground.Thispinalsoservesasthe negativeterminalofthecurrentsenseresistor.TheExposed Padmustbesoldereddirectlytothelocalgroundplane.
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LT3757 block DiagraM
L1 VIN R4 R3 CDC D1 VOUT L2 R2 + R1 COUT2 COUT1
*
+
CIN
*
9 A10 SHDN/UVLO 10 VIN
FBX
2.5V IS3 VC 1 CC2 CC1 RC 1.72V
IS1 2A 2.5V IS2 10A Q3 G4
- +
1.22V
INTERNAL REGULATOR AND UVLO UVLO G3
+ A9 -
A8
17.5V
CURRENT LIMIT 7.2V LDO INTVCC 8 CVCC
- + - +
A11
TSD 165C G6 VC
+ -
2.7V UP 2.6V DOWN DRIVER SR1 GATE O G2 7 M1
-0.88V
Q2 1.6V FBX FBX 2 -0.8V
PWM COMPARATOR A6 SLOPE RAMP VISENSE
+ A1 - + A2 -
FREQUENCY FOLDBACK
1.25V FREQ FOLDBACK
SS 3 CSS 5
Figure 1. LT3757 Block Diagram Working as a SEPIC Converter
+ -
1.25V
A3
SYNC 4
+ -
A12
A7
G5
R
S
- +
108mV SENSE 6 GND 11 RSENSE
RAMP GENERATOR G1 100kHz-1MHz OSCILLATOR
+ A5 -
+ + -
A4 FREQ PROG
Q1
RT
3757 F01
RT
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LT3757 applicaTions inForMaTion
Main Control Loop TheLT3757usesafixedfrequency,currentmodecontrol schemetoprovideexcellentlineandloadregulation.OperationcanbebestunderstoodbyreferringtotheBlock DiagraminFigure1. ThestartofeachoscillatorcyclesetstheSRlatch(SR1)and turnsontheexternalpowerMOSFETswitchM1through driverG2.Theswitchcurrentflowsthroughtheexternal currentsensingresistorRSENSEandgeneratesavoltage proportional to the switch current. This current sense voltageVISENSE(amplifiedbyA5)isaddedtoastabilizing slopecompensationrampandtheresultingsum(SLOPE) isfedintothepositiveterminalofthePWMcomparatorA7. WhenSLOPEexceedsthelevelatthenegativeinputofA7 (VCpin),SR1isreset,turningoffthepowerswitch.The levelatthenegativeinputofA7issetbytheerroramplifier A1(orA2)andisanamplifiedversionofthedifference betweenthefeedbackvoltage(FBXpin)andthereference voltage(1.6Vor-0.8V,dependingontheconfiguration). Inthismanner,theerroramplifiersetsthecorrectpeak switchcurrentleveltokeeptheoutputinregulation. TheLT3757hasaswitchcurrentlimitfunction.Thecurrent sensevoltageisinputtothecurrentlimitcomparatorA6. IftheSENSEpinvoltageishigherthanthesensecurrent limitthresholdVSENSE(MAX)(110mV,typical),A6willreset SR1andturnoffM1immediately. The LT3757 is capable of generating either positive or negativeoutputvoltagewithasingleFBXpin.Itcanbe configuredasaboost,flybackorSEPICconvertertogeneratepositiveoutputvoltage,orasaninvertingconverter togeneratenegativeoutputvoltage.Whenconfiguredas aSEPICconverter,asshowninFigure1,theFBXpinis pulleduptotheinternalbiasvoltageof1.6Vbyavoltagedivider(R1andR2)connectedfromVOUTtoGND. Comparator A2 becomes inactive and comparator A1 performstheinvertingamplificationfromFBXtoVC.When theLT3757isinaninvertingconfiguration,theFBXpin ispulleddownto-0.8Vbyavoltagedividerconnected fromVOUTtoGND.ComparatorA1becomesinactiveand comparatorA2performsthenoninvertingamplification fromFBXtoVC. The LT3757 has overvoltage protection functions to protect the converter from excessive output voltage overshootduringstart-uporrecoveryfromashort-circuit condition.AnovervoltagecomparatorA11(with20mV hysteresis)senseswhentheFBXpinvoltageexceedsthe positiveregulatedvoltage(1.6V)by8%andprovidesa reset pulse. Similarly, an overvoltage comparator A12 (with10mVhysteresis)senseswhentheFBXpinvoltage exceedsthenegativeregulatedvoltage(-0.8V)by11% andprovidesaresetpulse.Bothresetpulsesaresentto themainRSlatch(SR1)throughG6andG5.Thepower MOSFETswitchM1isactivelyheldoffforthedurationof anoutputovervoltagecondition. Programming Turn-On and Turn-Off Thresholds with the SHDN/UVLO Pin The SHDN/UVLO pin controls whether the LT3757 is enabled or is in shutdown state. A micropower 1.22V reference,acomparatorA10andacontrollablecurrent sourceIS1allowtheusertoaccuratelyprogramthesupplyvoltageatwhichtheICturnsonandoff.Thefalling valuecanbeaccuratelysetbytheresistordividersR3 andR4.WhenSHDN/UVLOisabove0.7V,andbelowthe 1.22Vthreshold,thesmallpull-downcurrentsourceIS1 (typical2A)isactive. Thepurposeofthiscurrentistoallowtheusertoprogram therisinghysteresis.TheBlockDiagramofthecomparator andtheexternalresistorsisshowninFigure1.Thetypical fallingthresholdvoltageandrisingthresholdvoltagecan becalculatedbythefollowingequations: (R3 + R4) R4 VVIN, RISING = 2A * R3+ VIN, FALLING VVIN, FALLING = 1.22 *
3757fb
LT3757 applicaTions inForMaTion
ForapplicationswheretheSHDN/UVLOpinisonlyused asalogicinput,theSHDN/UVLOpincanbeconnected directlytotheinputvoltageVINforalways-onoperation. INTVCC Regulator Bypassing and Operation Aninternal,lowdropout(LDO)voltageregulatorproduces the7.2VINTVCCsupplywhichpowersthegatedriver,as showninFigure1.Ifalowinputvoltageoperationisexpected(e.g.,supplyingpowerfromalithium-ionbattery ora3.3Vlogicsupply),lowthresholdMOSFETsshould beused.TheLT3757containsanundervoltagelockout comparatorA8andanovervoltagelockoutcomparator A9fortheINTVCCsupply.TheINTVCCundervoltage(UV) threshold is 2.7V (typical), with 100mV hysteresis, to ensurethattheMOSFETshavesufficientgatedrivevoltage beforeturningon.ThelogiccircuitrywithintheLT3757is alsopoweredfromtheinternalINTVCCsupply. TheINTVCCovervoltage(OV)thresholdissettobe17.5V (typical)toprotectthegateofthepowerMOSFET.When INTVCCisbelowtheUVthreshold,orabovetheOVthreshold,theGATEpinwillbeforcedtoGNDandthesoft-start operationwillbetriggered. TheINTVCCregulatormustbebypassedtogroundimmediatelyadjacenttotheICpinswithaminimumof4.7F ceramiccapacitor.Goodbypassingisnecessarytosupply thehightransientcurrentsrequiredbytheMOSFETgate driver. Inanactualapplication,mostoftheICsupplycurrentis usedtodrivethegatecapacitanceofthepowerMOSFET. Theon-chippowerdissipationcanbeasignificantconcern when a large power MOSFET is being driven at a high frequencyandtheVINvoltageishigh.Itisimportantto limitthepowerdissipationthroughselectionofMOSFET and/oroperatingfrequencysotheLT3757doesnotexceed its maximum junction temperature rating. The junction temperature TJ can be estimated using the following equations: TJ=TA+PIC*JA TA=ambienttemperature JA=junction-to-ambientthermalresistance PIC=ICpowerconsumption =VIN*(IQ+IDRIVE) IQ=VINoperationIQ=1.6mA IDRIVE=averagegatedrivecurrent=f*QG f=switchingfrequency QG=powerMOSFETtotalgatecharge TheLT3757usespackageswithanExposedPadforenhancedthermalconduction.Withpropersolderingtothe ExposedPadontheundersideofthepackageandafull copperplaneunderneaththedevice,thermalresistance (JA)willbeabout43C/WfortheDDpackageand40C/W fortheMSEpackage.Foranambientboardtemperatureof TA=70Candmaximumjunctiontemperatureof125C, themaximumIDRIVE(IDRIVE(MAX))oftheDDpackagecan becalculatedas: IDRIVE(MAX ) = (TJ - TA ) 1.28 W - IQ = - 1.6mA VIN ( JA * VIN )
TheLT3757hasaninternalINTVCCIDRIVEcurrentlimit functiontoprotecttheICfromexcessiveon-chippower dissipation.TheIDRIVEcurrentlimitdecreasesastheVIN increases(seetheINTVCCMinimumOutputCurrentvsVIN graphintheTypicalPerformanceCharacteristicssection). IfIDRIVEreachesthecurrentlimit,INTVCCvoltagewillfall andmaytriggerthesoft-start. BasedontheprecedingequationandtheINTVCCMinimum OutputCurrentvsVINgraph,theusercancalculatethe maximumMOSFETgatechargetheLT3757candriveat agivenVINandswitchfrequency.Aplotofthemaximum QGvsVINatdifferentfrequenciestoguaranteeaminimum 4.5VINTVCCisshowninFigure2. AsillustratedinFigure2,atrade-offbetweentheoperating frequencyandthesizeofthepowerMOSFETmaybeneeded inordertomaintainareliableICjunctiontemperature. Prior to lowering the operating frequency, however, be suretocheckwithpowerMOSFETmanufacturersfortheir mostrecentlowQG,lowRDS(ON)devices.PowerMOSFET manufacturingtechnologiesarecontinuallyimproving,with newerandbetterperformancedevicesbeingintroduced almostyearly.
3757fb
0
LT3757 applicaTions inForMaTion
300 250 300kHz 200 QG (nC) 150 100 1MHz 50 0 GND
3757 F03
LT3757 INTVCC
DVCC
RVCC
VOUT
CVCC 4.7F
Figure 3. Connecting INTVCC to VOUT
0
5
10
15
20 VIN (V)
25
30
35
40
3757 F02
ornottheINTVCCpinisconnectedtoanexternalvoltage source,itisalwaysnecessarytohavethedrivercircuitry bypassedwitha4.7FlowESRceramiccapacitortoground immediatelyadjacenttotheINTVCCandGNDpins. Operating Frequency and Synchronization The choice of operating frequency may be determined byon-chippowerdissipation,otherwiseitisatrade-off betweenefficiencyandcomponentsize.Lowfrequency operationimprovesefficiencybyreducinggatedrivecurrentandMOSFETanddiodeswitchinglosses.However, lower frequency operation requires a physically larger inductor.Switchingfrequencyalsohasimplicationsfor loopcompensation.TheLT3757usesaconstant-frequency architecturethatcanbeprogrammedovera100kHzto 1000kHzrangewithasingleexternalresistorfromthe RTpintoground,asshowninFigure1.TheRTpinmust haveanexternalresistortoGNDforproperoperationof theLT3757.AtableforselectingthevalueofRTforagiven operatingfrequencyisshowninTable1.
Table 1. Timing Resistor (RT ) Value
OSCILLATOR FREQUENCY (kHz) 100 200 300 400 500 600 700 800 900 1000 RT (k) 140 63.4 41.2 30.9 24.3 19.6 16.5 14 12.1 10.5
Figure 2. Recommended Maximum QG vs VIN at Different Frequencies to Ensure INTVCC Higher Than 4.5V
Aneffectiveapproachtoreducethepowerconsumption oftheinternalLDOforgatedriveistotietheINTVCCpin toanexternalvoltagesourcehighenoughtoturnoffthe internalLDOregulator. If the input voltage VIN does not exceed the absolute maximumratingofboththepowerMOSFETgate-source voltage(VGS)andtheINTVCCovervoltagelockoutthreshold voltage(17.5V),theINTVCCpincanbeshorteddirectly totheVINpin.Inthiscondition,theinternalLDOwillbe turned off and the gate driver will be powered directly fromtheinputvoltage,VIN.WiththeINTVCCpinshortedto VIN,however,asmallcurrent(around16A)willloadthe INTVCCinshutdownmode.Forapplicationsthatrequire thelowestshutdownmodeinputsupplycurrent,donot connecttheINTVCCpintoVIN. InSEPICorflybackapplications,theINTVCCpincanbe connectedtotheoutputvoltageVOUTthroughablocking diode,asshowninFigure3,ifVOUTmeetsthefollowing conditions: 1.VOUT3757fb
LT3757 applicaTions inForMaTion
TheoperatingfrequencyoftheLT3757canbesynchronized toanexternalclocksource.Byprovidingadigitalclock signalintotheSYNCpin,theLT3757willoperateatthe SYNCclockfrequency.Ifthisfeatureisused,anRTresistor shouldbechosentoprogramaswitchingfrequency20% slowerthanSYNCpulsefrequency.TheSYNCpulseshould haveaminimumpulsewidthof200ns.TietheSYNCpin toGNDifthisfeatureisnotused. Duty Cycle Consideration Switchingdutycycleisakeyvariabledefiningconverter operation.Assuch,itslimitsmustbeconsidered.Minimum on-timeisthesmallesttimedurationthattheLT3757is capable of turning on the power MOSFET. This time is generallyabout220ns(typical)(seeMinimumOn-Time intheElectricalCharacteristicstable).Ineachswitching cycle,theLT3757keepsthepowerswitchoffforatleast 220ns(typical)(seeMinimumOff-TimeintheElectrical Characteristicstable). The minimum on-time and minimum off-time and the switchingfrequencydefinetheminimumandmaximum switchingdutycyclesaconverterisabletogenerate: Minimumdutycycle=minimumon-time*frequency Maximumdutycycle=1-(minimumoff-time*frequency) Programming the Output Voltage Theoutputvoltage(VOUT)issetbyaresistordivider,as showninFigure1.ThepositiveandnegativeVOUTareset bythefollowingequations: R2 VOUT, POSITIVE = 1.6 V * 1+ R1 R2 VOUT, NEGATIVE = -0.8 V * 1+ R1 The resistors R1 and R2 are typically chosen so that theerrorcausedbythecurrentflowingintotheFBXpin duringnormaloperationislessthan1%(thistranslates toamaximumvalueofR1atabout158k). Soft-Start TheLT3757containsseveralfeaturestolimitpeakswitch currents and output voltage (VOUT) overshoot during start-uporrecoveryfromafaultcondition.Theprimary purposeofthesefeaturesistopreventdamagetoexternal componentsortheload. Highpeakswitchcurrentsduringstart-upmayoccurin switchingregulators.SinceVOUTisfarfromitsfinalvalue, thefeedbackloopissaturatedandtheregulatortriesto chargetheoutputcapacitorasquicklyaspossible,resulting inlargepeakcurrents.Alargesurgecurrentmaycause inductorsaturationorpowerswitchfailure. TheLT3757addressesthismechanismwiththeSSpin.As showninFigure1,theSSpinreducesthepowerMOSFET currentbypullingdowntheVCpinthroughQ2.Inthisway theSSallowstheoutputcapacitortochargegraduallytowarditsfinalvaluewhilelimitingthestart-uppeakcurrents. Thetypicalstart-upwaveformsareshownintheTypical PerformanceCharacteristicssection.Theinductorcurrent ILslewingrateislimitedbythesoft-startfunction. Besidesstart-up,soft-startcanalsobetriggeredbythe followingfaults: 1.INTVCC>17.5V 2.INTVCC<2.6V 3.Thermallockout AnyofthesethreefaultswillcausetheLT3757tostop switchingimmediately.TheSSpinwillbedischargedby Q3.WhenallfaultsareclearedandtheSSpinhasbeen dischargedbelow0.2V,a10AcurrentsourceIS2starts chargingtheSSpin,initiatingasoft-startoperation. The soft-start interval is set by the soft-start capacitor selectionaccordingtotheequation: TSS = CSS * 1.25V 10A
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FBX Frequency Foldback WhenVOUTisverylowduringstart-uporashort-circuit faultontheoutput,theswitchingregulatormustoperate atlowdutycyclestomaintainthepowerswitchcurrent withinthecurrentlimitrange,sincetheinductorcurrent decayrateisverylowduringswitchofftime.Theminimum on-timelimitationmaypreventtheswitcherfromattaining asufficientlylowdutycycleattheprogrammedswitchingfrequency.So,theswitchcurrentwillkeepincreasing througheachswitchcycle,exceedingtheprogrammed currentlimit.Topreventtheswitchpeakcurrentsfrom exceedingtheprogrammedvalue,theLT3757contains a frequency foldback function to reduce the switching frequencywhentheFBXvoltageislow(seetheNormalized Switching Frequency vs FBX graph in the Typical PerformanceCharacteristicssection). Thetypicalfrequencyfoldbackwaveformsareshownin theTypicalPerformanceCharacteristicssection.ThefrequencyfoldbackfunctionpreventsILfromexceedingthe programmedlimitsbecauseoftheminimumon-time. Duringfrequencyfoldback,externalclocksynchronization is disabled to prevent interference with frequency reducingoperation. Thermal Lockout If LT3757 die temperature reaches 165C (typical), the partwillgointothermallockout.Thepowerswitchwill beturnedoff.Asoft-startoperationwillbetriggered.The partwillbeenabledagainwhenthedietemperaturehas droppedby5C(nominal). Loop Compensation Loopcompensationdeterminesthestabilityandtransient performance.TheLT3757usescurrentmodecontrolto regulatetheoutputwhichsimplifiesloopcompensation. Theoptimumvaluesdependontheconvertertopology,the componentvaluesandtheoperatingconditions(including theinputvoltage,loadcurrent,etc.).Tocompensatethe feedbackloopoftheLT3757,aseriesresistor-capacitor networkisusuallyconnectedfromtheVCpintoGND. Figure1showsthetypicalVCcompensationnetwork.For mostapplications,thecapacitorshouldbeintherangeof 470pFto22nF ,andtheresistorshouldbeintherangeof 5kto50k.AsmallcapacitorisoftenconnectedinparallelwiththeRCcompensationnetworktoattenuatethe VCvoltagerippleinducedfromtheoutputvoltageripple throughtheinternalerroramplifier.Theparallelcapacitor usuallyrangesinvaluefrom10pFto100pF .Apractical approachtodesignthecompensationnetworkistostart withoneofthecircuitsinthisdatasheetthatissimilar toyourapplication,andtunethecompensationnetwork to optimize the performance. Stability should then be checkedacrossalloperatingconditions,includingload current,inputvoltageandtemperature. SENSE Pin Programming For control and protection, the LT3757 measures the powerMOSFETcurrentbyusingasenseresistor(RSENSE) betweenGNDandtheMOSFETsource.Figure4showsa typicalwaveformofthesensevoltage(VSENSE)acrossthe senseresistor.ItisimportanttouseKelvintracesbetween theSENSEpinandRSENSE,andtoplacetheICGNDas closeaspossibletotheGNDterminaloftheRSENSEfor properoperation.
VSENSE VSENSE = VSENSE(MAX) VSENSE(PEAK) t TS
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VSENSE(MAX)
DTS
Figure 4. The Sense Voltage During a Switching Cycle
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DuetothecurrentlimitfunctionoftheSENSEpin,RSENSE shouldbeselectedtoguaranteethatthepeakcurrentsense voltageVSENSE(PEAK)duringsteadystatenormaloperation islowerthantheSENSEcurrentlimitthreshold(seethe Electrical Characteristics table). Given a 20% margin, VSENSE(PEAK) is set to be 80mV. Then, the maximum switchripplecurrentpercentagecanbecalculatedusing thefollowingequation: cisusedinsubsequentdesignexamplestocalculateinductorvalue.VSENSEistheripplevoltageacrossRSENSE. TheLT3757switchingcontrollerincorporates100nstiming intervaltoblanktheringingonthecurrentsensesignal immediatelyafterM1isturnedon.Thisringingiscaused bytheparasiticinductanceandcapacitanceofthePCB trace,thesenseresistor,thediode,andtheMOSFET.The 100nstimingintervalisadequateformostoftheLT3757 applications.Intheapplicationsthathaveverylargeand longringingonthecurrentsensesignal,asmallRCfilter can be added to filter out the excess ringing. Figure 5 showstheRCfilteronSENSEpin.Itisusuallysufficient to choose 22 for RFLT and 2.2nF to 10nF for CFLT. KeepRFLT'sresistancelow.Rememberthatthereis65A (typical)flowingoutoftheSENSEpin.AddingRFLTwill affecttheSENSEcurrentlimitthreshold: VSENSE_ILIM=108mV-65A*RFLT
GATE LT3757 SENSE GND CFLT RSENSE RFLT M1
APPLICATION CIRCUITS TheLT3757canbeconfiguredasdifferenttopologies.The firsttopologytobeanalyzedwillbetheboostconverter, followedbytheflyback,SEPICandinvertingconverters. Boost Converter: Switch Duty Cycle and Frequency TheLT3757canbeconfiguredasaboostconverterfor the applications where the converter output voltage is higherthantheinputvoltage.Rememberthatboostconvertersarenotshort-circuitprotected.Underashorted outputcondition,theinductorcurrentislimitedonlyby theinputsupplycapability.Forapplicationsrequiringa step-upconverterthatisshort-circuitprotected,please refer to the Applications Information section covering SEPICconverters. Theconversionratioasafunctionofdutycycleis VOUT 1 = VIN 1- D
c=
VSENSE 80mV - 0.5 * VSENSE
incontinuousconductionmode(CCM). ForaboostconverteroperatinginCCM,thedutycycle ofthemainswitchcanbecalculatedbasedontheoutput voltage(VOUT)andtheinputvoltage(VIN).Themaximum duty cycle (DMAX) occurs when the converter has the minimuminputvoltage: DMAX = VOUT - VIN(MIN) VOUT
Discontinuousconductionmode(DCM)provideshigher conversionratiosatagivenfrequencyatthecostofreduced efficienciesandhigherswitchingcurrents.
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Figure 5. The RC Filter on SENSE Pin
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Boost Converter: Inductor and Sense Resistor Selection Fortheboosttopology,themaximumaverageinductor currentis: 1 IL(MAX ) = IO(MAX ) * 1- DMAX Then,theripplecurrentcanbecalculatedby: IL = c * IL(MAX ) = c * IO(MAX ) * 1 1- DMAX Based on these equations, the user should choose the inductorshavingsufficientsaturationandRMScurrent ratings. SetthesensevoltageatIL(PEAK)tobetheminimumofthe SENSEcurrentlimitthresholdwitha20%margin.The senseresistorvaluecanthenbecalculatedtobe: RSENSE = 80 mV IL(PEAK )
Theconstantcintheprecedingequationrepresentsthe percentage peak-to-peak ripple current in the inductor, relativetoIL(MAX). Theinductorripplecurrenthasadirecteffectonthechoice of the inductor value. Choosing smaller values of IL requireslargeinductancesandreducesthecurrentloop gain(theconverterwillapproachvoltagemode).Accepting largervaluesofILprovidesfasttransientresponseand allowstheuseoflowinductances,butresultsinhigherinput currentrippleandgreatercorelosses.Itisrecommended thatcfallwithintherangeof0.2to0.6. Givenanoperatinginputvoltagerange,andhavingchosen theoperatingfrequencyandripplecurrentintheinductor, theinductorvalueoftheboostconvertercanbedetermined usingthefollowingequation: L= VIN(MIN) IL * f * DMAX
Boost Converter: Power MOSFET Selection ImportantparametersforthepowerMOSFETincludethe drain-sourcevoltagerating(VDS),thethresholdvoltage (VGS(TH)),theon-resistance(RDS(ON)),thegatetosource andgatetodraincharges(QGSandQGD),themaximum drain current (ID(MAX)) and the MOSFET's thermal resistances(RJCandRJA). ThepowerMOSFETwillseefulloutputvoltage,plusa diodeforwardvoltage,andanyadditionalringingacross itsdrain-to-sourceduringitsoff-time.Itisrecommended tochooseaMOSFETwhoseBVDSSishigherthanVOUTby asafetymargin(a10Vsafetymarginisusuallysufficient). ThepowerdissipatedbytheMOSFETinaboostconverteris: PFET=I2L(MAX)*RDS(ON)*DMAX+2*V2OUT*IL(MAX) *CRSS*f/1A Thefirsttermintheprecedingequationrepresentsthe conductionlossesinthedevice,andthesecondterm,the switchingloss.CRSSisthereversetransfercapacitance, whichisusuallyspecifiedintheMOSFETcharacteristics. For maximum efficiency, RDS(ON) and CRSS should be minimized.Fromaknownpowerdissipatedinthepower MOSFET,itsjunctiontemperaturecanbeobtainedusing thefollowingequation: TJ=TA+PFET*JA=TA+PFET*(JC+CA)
ThepeakandRMSinductorcurrentare: c IL(PEAK ) = IL(MAX ) * 1+ 2 IL(RMS) = IL(MAX ) * 1+ c2 12
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TJ must not exceed the MOSFET maximum junction temperature rating. It is recommended to measure the MOSFETtemperatureinsteadystatetoensurethatabsolute maximumratingsarenotexceeded. Boost Converter: Output Diode Selection Tomaximizeefficiency,afastswitchingdiodewithlow forwarddropandlowreverseleakageisdesirable.The peak reverse voltage that the diode must withstand is equaltotheregulatoroutputvoltageplusanyadditional ringingacrossitsanode-to-cathodeduringtheon-time. Theaverageforwardcurrentinnormaloperationisequal totheoutputcurrent,andthepeakcurrentisequalto: c ID(PEAK ) = IL(PEAK ) = 1+ * IL(MAX ) 2 Itisrecommendedthatthepeakrepetitivereversevoltage ratingVRRMishigherthanVOUTbyasafetymargin(a10V safetymarginisusuallysufficient). Thepowerdissipatedbythediodeis: PD=IO(MAX)*VD andthediodejunctiontemperatureis: TJ=TA+PD*RJA TheRJAtobeusedinthisequationnormallyincludes theRJCforthedeviceplusthethermalresistancefrom theboardtotheambienttemperatureintheenclosure.TJ mustnotexceedthediodemaximumjunctiontemperature rating. Boost Converter: Output Capacitor Selection ContributionsofESR(equivalentseriesresistance),ESL (equivalentseriesinductance)andthebulkcapacitance must be considered when choosing the correct output capacitorsforagivenoutputripplevoltage.Theeffectof thesethreeparameters(ESR,ESLandbulkC)ontheoutput voltageripplewaveformforatypicalboostconverteris illustratedinFigure6.
tON tOFF VCOUT VOUT (AC) VESR
RINGING DUE TO TOTAL INDUCTANCE (BOARD + CAP)
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Figure 6. The Output Ripple Waveform of a Boost Converter
Thechoiceofcomponent(s)beginswiththemaximum acceptableripplevoltage(expressedasapercentageof theoutputvoltage),andhowthisrippleshouldbedivided betweentheESRstepVESRandthecharging/dischargingVCOUT.Forthepurposeofsimplicity,wewillchoose 2%forthemaximumoutputripple,tobedividedequally betweenVESRandVCOUT.Thispercentageripplewill change,dependingontherequirementsoftheapplication,andthefollowingequationscaneasilybemodified. Fora1%contributiontothetotalripplevoltage,theESR oftheoutputcapacitorcanbedeterminedusingthefollowingequation: ESRCOUT 0.01* VOUT ID(PEAK )
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ForthebulkCcomponent,whichalsocontributes1%to thetotalripple: COUT IO(MAX ) 0.01* VOUT * f FLYBACK CONVERTER APPLICATIONS TheLT3757canbeconfiguredasaflybackconverterforthe applicationswheretheconvertershavemultipleoutputs, highoutputvoltagesorisolatedoutputs.Figure7shows asimplifiedflybackconverter. Theflybackconverterhasaverylowpartscountformultipleoutputs,andwithprudentselectionofturnsratio,can havehighoutput/inputvoltageconversionratioswitha desirabledutycycle.However,ithaslowefficiencydueto thehighpeakcurrents,highpeakvoltagesandconsequent powerloss.Theflybackconverteriscommonlyusedfor anoutputpoweroflessthan50W. Theflybackconvertercanbedesignedtooperateeither incontinuousordiscontinuousmode.Comparedtocontinuousmode,discontinuousmodehastheadvantageof smallertransformerinductancesandeasyloopcompensation,andthedisadvantageofhigherpeak-to-average currentandlowerefficiency.Inthehighoutputvoltage applications, the flyback converters can be designed to operate in discontinuous mode to avoid using large transformers.
VIN SUGGESTED RCD SNUBBER NP:NS D
Theoutputcapacitorinaboostregulatorexperienceshigh RMSripplecurrents,asshowninFigure6.TheRMSripple currentratingoftheoutputcapacitorcanbedetermined usingthefollowingequation: IRMS(COUT ) IO(MAX ) * DMAX 1- DMAX
MultiplecapacitorsareoftenparalleledtomeetESRrequirements.Typically,oncetheESRrequirementissatisfied,the capacitanceisadequateforfilteringandhastherequired RMScurrentrating.Additionalceramiccapacitorsinparallelarecommonlyusedtoreducetheeffectofparasitic inductanceintheoutputcapacitor,whichreduceshigh frequencyswitchingnoiseontheconverteroutput. Boost Converter: Input Capacitor Selection Theinputcapacitorofaboostconverterislesscritical thantheoutputcapacitor,duetothefactthattheinductor isinserieswiththeinput,andtheinputcurrentwaveformiscontinuous.Theinputvoltagesourceimpedance determinesthesizeoftheinputcapacitor,whichistypicallyintherangeof10Fto100F .AlowESRcapacitor isrecommended,althoughitisnotascriticalasforthe outputcapacitor. TheRMSinputcapacitorripplecurrentforaboostconverteris: IRMS(CIN)=0.3*IL
+
CIN
VSN
- +
CSN RSN LP LS ID
+
+
COUT
DSN ISW LT3757 GATE SENSE M
-
+ -
VDS
RSENSE GND
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Figure 7. A Simplified Flyback Converter
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Flyback Converter: Switch Duty Cycle and Turns Ratio Theflybackconverterconversionratiointhecontinuous modeoperationis: VOUT NS D = * VIN NP 1- D Accordingtotheprecedingequations,theuserhasrelative freedominselectingtheswitchdutycycleorturnsratioto suitagivenapplication.Theselectionsofthedutycycle andtheturnsratioaresomewhatiterativeprocesses,due tothenumberofvariablesinvolved.Theusercanchoose eitheradutycycleoraturnsratioasthestartpoint.The followingtrade-offsshouldbeconsideredwhenselectingtheswitchdutycycleorturnsratio,tooptimizethe converter performance. A higher duty cycle affects the flybackconverterinthefollowingaspects: * Lower MOSFET RMS current ISW(RMS), but higher MOSFETVDSpeakvoltage * Lower diode peak reverse voltage, but higher diode RMScurrentID(RMS) * Highertransformerturnsratio(NP/NS) Thechoice, D 1 = D + D2 3 (fordiscontinuousmodeoperationwithagivenD3)gives thepowerMOSFETthelowestpowerstress(theproduct ofRMScurrentandpeakvoltage).However,inthehigh outputvoltageapplications,ahigherdutycyclemaybe adopted to limit the large peak reverse voltage of the diode.Thechoice,
ISW(MAX)
whereNS/NPisthesecondtoprimaryturnsratio. Figure8showsthewaveformsoftheflybackconverter indiscontinuousmodeoperation.Duringeachswitching period TS, three subintervals occur: DTS, D2TS, D3TS. During DTS, M is on, and D is reverse-biased. During D2TS,Misoff,andLSisconductingcurrent.BothLPand LScurrentsarezeroduringD3TS. Theflybackconverterconversionratiointhediscontinuousmodeoperationis: VOUT NS D = * VIN NP D2
VDS
ISW
D 2 = D + D2 3 (fordiscontinuousmodeoperationwithagivenD3)gives thediodethelowestpowerstress(theproductofRMS currentandpeakvoltage).Anextremehighorlowduty cycleresultsinhighpowerstressontheMOSFETordiode, andreducesefficiency.Itisrecommendedtochoosea dutycycle,D,between20%and80%.
ID
ID(MAX) DTS D2TS TS D3TS t
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Figure 8. Waveforms of the Flyback Converter in Discontinuous Mode Operation
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Flyback Converter: Transformer Design for Discontinuous Mode Operation Thetransformerdesignfordiscontinuousmodeofoperationischosenaspresentedhere.AccordingtoFigure8, the minimum D3 (D3MIN) occurs when the converter hastheminimumVINandthemaximumoutputpower (POUT).ChooseD3MINtobeequaltoorhigherthan10% toguaranteetheconverterisalwaysindiscontinuous mode operation (choosing higher D3 allows the use oflowinductances,butresultsinahigherswitchpeak current). TheusercanchooseaDMAXasthestartpoint.Then,the maximumaverageprimarycurrentscanbecalculatedby thefollowingequation: ILP(MAX ) = ISW(MAX ) = POUT(MAX ) DMAX * VIN(MIN) * h AccordingtoFigure8,theprimaryandsecondarypeak currentsare: ILP(PEAK)=ISW(PEAK)=2*ILP(MAX) ILS(PEAK)=ID(PEAK)=2*ILS(MAX) Theprimaryandsecondinductorvaluesoftheflyback convertertransformercanbedeterminedusingthefollowingequations: LP = D2MAX * V 2IN(MAX ) * h 2 * POUT(MAX ) * f
D22 *( VOUT + VD) LS = 2 * I OUT(MAX ) * f Theprimarytosecondturnsratiois: NP L =P LS NS Flyback Converter: Snubber Design Transformer leakage inductance (on either the primary orsecondary)causesavoltagespiketooccurafterthe MOSFETturn-off.Thisisincreasinglyprominentathigher load currents, where more stored energy must be dissipated.Insomecasesasnubbercircuitwillberequired toavoidovervoltagebreakdownattheMOSFET'sdrain node.Therearedifferentsnubbercircuits,andApplication Note19isagoodreferenceonsnubberdesign.AnRCD snubberisshowninFigure7. Thesnubberresistorvalue(RSN)canbecalculatedbythe followingequation: V 2SN - VSN * VOUT * NP NS
wherehistheconverterefficiency. Iftheflybackconverterhasmultipleoutputs,POUT(MAX) isthesumofalltheoutputpower. Themaximumaveragesecondarycurrentis: ILS(MAX ) = ID(MAX ) = where: D2=1-DMAX-D3 theprimaryandsecondaryRMScurrentsare: ILP(RMS) = 2 * ILP(MAX ) * ILS(RMS) = 2 * ILS(MAX ) * DMAX 3 D2 3 IOUT(MAX ) D2
RSN = 2 *
I2SW(PEAK ) * L LK * f
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where VSN is the snubber capacitor voltage. A smaller VSNresultsinalargersnubberloss.AreasonableVSNis 2to2.5timesof: VOUT * NP NS Flyback Converter: Power MOSFET Selection Fortheflybackconfiguration,theMOSFETisselectedwith aVDCratinghighenoughtohandlethemaximumVIN,the reflectedsecondaryvoltageandthevoltagespikedueto theleakageinductance.ApproximatetherequiredMOSFET VDCratingusing: BVDSS>VDS(PEAK) where:
VDS(PEAK ) = VIN(MAX ) + VSN
LLK is the leakage inductance of the primary winding, whichisusuallyspecifiedinthetransformercharacteristics. LLK can be obtained by measuring the primary inductance with the secondary windings shorted. The snubbercapacitorvalue(CCN)canbedeterminedusing thefollowingequation: CCN = VSN VSN * RCN * f
ThepowerdissipatedbytheMOSFETinaflybackconverteris: PFET=I2M(RMS)*RDS(ON)+2*V2DS(PEAK)*IL(MAX)* CRSS*f/1A Thefirstterminthisequationrepresentstheconduction lossesinthedevice,andthesecondterm,theswitching loss.CRSSisthereversetransfercapacitance,whichis usuallyspecifiedintheMOSFETcharacteristics. FromaknownpowerdissipatedinthepowerMOSFET,its junctiontemperaturecanbeobtainedusingthefollowing equation: TJ=TA+PFET*JA=TA+PFET*(JC+CA) TJ must not exceed the MOSFET maximum junction temperature rating. It is recommended to measure the MOSFETtemperatureinsteadystatetoensurethatabsolute maximumratingsarenotexceeded.
whereVSNisthevoltagerippleacrossCCN.Areasonable VSNis5%to10%ofVSN.Thereversevoltageratingof DSNshouldbehigherthanthesumofVSNandVIN(MAX). Flyback Converter: Sense Resistor Selection Inaflybackconverter,whenthepowerswitchisturned on, the current flowing through the sense resistor (ISENSE)is: ISENSE=ILP SetthesensevoltageatILP(PEAK)tobetheminimumof theSENSEcurrentlimitthresholdwitha20%margin.The senseresistorvaluecanthenbecalculatedtobe: RSENSE = 80 mV ILP(PEAK )
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Flyback Converter: Output Diode Selection Theoutputdiodeinaflybackconverterissubjecttolarge RMScurrentandpeakreversevoltagestresses.Afast switchingdiodewithalowforwarddropandalowreverse leakageisdesired.Schottkydiodesarerecommendedif theoutputvoltageisbelow100V. Approximatetherequiredpeakrepetitivereversevoltage ratingVRRMusing: N VRRM > S * VIN(MAX ) + VOUT NP Flyback Converter: Input Capacitor Selection Theinputcapacitorinaflybackconverterissubjectto a large RMS current due to the discontinuous primary current. To prevent large voltage transients, use a low ESRinputcapacitorsizedforthemaximumRMScurrent. TheRMSripplecurrentratingoftheinputcapacitorsin discontinuous operation can be determined using the followingequation:
IRMS(CIN),DISCONTINUOUS POUT(MAX ) VIN(MIN) * h * 4 - (3 * DMAX ) 3 * DMAX
Thepowerdissipatedbythediodeis: PD=IO(MAX)*VD andthediodejunctiontemperatureis: TJ=TA+PD*RJA TheRJAtobeusedinthisequationnormallyincludes theRJCforthedevice,plusthethermalresistancefrom theboardtotheambienttemperatureintheenclosure.TJ mustnotexceedthediodemaximumjunctiontemperature rating. Flyback Converter: Output Capacitor Selection Theoutputcapacitoroftheflybackconverterhasasimilar operationconditionasthatoftheboostconverter.Referto theBoostConverter:OutputCapacitorSelectionsection forthecalculationofCOUTandESRCOUT. TheRMSripplecurrentratingoftheoutputcapacitors indiscontinuousoperationcanbedeterminedusingthe followingequation:
IRMS(COUT ),DISCONTINUOUS IO(MAX ) * 4 - (3 * D2) 3 * D2
SEPIC CONVERTER APPLICATIONS TheLT3757canbeconfiguredasaSEPIC(single-ended primaryinductanceconverter),asshowninFigure1.This topologyallowsfortheinputtobehigher,equal,orlower thanthedesiredoutputvoltage.Theconversionratioas afunctionofdutycycleis: VOUT + VD D = VIN 1- D
incontinuousconductionmode(CCM). InaSEPICconverter,noDCpathexistsbetweentheinput andoutput.Thisisanadvantageovertheboostconverter forapplicationsrequiringtheoutputtobedisconnected fromtheinputsourcewhenthecircuitisinshutdown. Comparedtotheflybackconverter,theSEPICconverter hastheadvantagethatboththepowerMOSFETandthe outputdiodevoltagesareclampedbythecapacitors(CIN, CDC and COUT), therefore, there is less voltage ringing across the power MOSFET and the output diodes. The SEPICconverterrequiresmuchsmallerinputcapacitors thanthoseoftheflybackconverter.Thisisduetothefact
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that,intheSEPICconverter,theinductorL1isinseries withtheinput,andtheripplecurrentflowingthroughthe inputcapacitoriscontinuous. SEPIC Converter: Switch Duty Cycle and Frequency ForaSEPICconverteroperatinginCCM,thedutycycle ofthemainswitchcanbecalculatedbasedontheoutput voltage (VOUT), the input voltage (VIN) and the diode forwardvoltage(VD). Themaximumdutycycle(DMAX)occurswhentheconverter hastheminimuminputvoltage: DMAX = VOUT + VD VIN(MIN) + VOUT + VD InaSEPICconverter,theswitchcurrentisequaltoIL1+ IL2whenthepowerswitchison,therefore,themaximum averageswitchcurrentisdefinedas: ISW(MAX ) = IL1(MAX ) + IL2(MAX ) = IO(MAX ) * 1 1- DMAX
andthepeakswitchcurrentis: c 1 ISW(PEAK ) = 1+ * IO(MAX ) * 1- DMAX 2
SEPIC Converter: Inductor and Sense Resistor Selection AsshowninFigure1,theSEPICconvertercontainstwo inductors:L1andL2.L1andL2canbeindependent,but canalsobewoundonthesamecore,sinceidenticalvoltagesareappliedtoL1andL2throughouttheswitching cycle. For the SEPIC topology, the current through L1 is the converterinputcurrent.Basedonthefactthat,ideally,the outputpowerisequaltotheinputpower,themaximum averageinductorcurrentsofL1andL2are: IL1(MAX ) = IIN(MAX ) = IO(MAX ) * IL 2(MAX ) =IO(MAX )
ISW
Theconstantcintheprecedingequationsrepresentsthe percentagepeak-to-peakripplecurrentintheswitch,relativetoISW(MAX),asshowninFigure9.Then,theswitch ripplecurrentISWcanbecalculatedby: ISW=c*ISW(MAX) TheinductorripplecurrentsIL1andIL2areidentical: IL1=IL2=0.5*ISW The inductor ripple current has a direct effect on the choiceoftheinductorvalue.Choosingsmallervaluesof ILrequireslargeinductancesandreducesthecurrent loop gain (the converter will approach voltage mode). AcceptinglargervaluesofILallowstheuseoflowinductances,butresultsinhigherinputcurrentrippleand greatercorelosses.Itisrecommendedthatcfallsinthe rangeof0.2to0.4.
DMAX 1- DMAX
ISW = ISW(MAX)
ISW(MAX)
DTS TS
t
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Figure 9. The Switch Current Waveform of the SEPIC Converter
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LT3757 applicaTions inForMaTion
Givenanoperatinginputvoltagerange,andhavingchosen theoperatingfrequencyandripplecurrentintheinductor,theinductorvalue(L1andL2areindependent)ofthe SEPICconvertercanbedeterminedusingthefollowing equation: L1= L2 = VIN(MIN) 0.5 * ISW * f * DMAX Basedontheprecedingequations,theusershouldchoose theinductorshavingsufficientsaturationandRMScurrentratings. InaSEPICconverter,whenthepowerswitchisturnedon, thecurrentflowingthroughthesenseresistor(ISENSE)is theswitchcurrent. SetthesensevoltageatISENSE(PEAK)tobetheminimum oftheSENSEcurrentlimitthresholdwitha20%margin. Thesenseresistorvaluecanthenbecalculatedtobe: RSENSE = 80 mV ISW(PEAK )
FormostSEPICapplications,theequalinductorvalues willfallintherangeof1Hto100H. BymakingL1=L2,andwindingthemonthesamecore,the valueofinductanceintheprecedingequationisreplaced by2L,duetomutualinductance: L= VIN(MIN) ISW * f * DMAX
SEPIC Converter: Power MOSFET Selection For the SEPIC configuration, choose a MOSFET with a VDCratinghigherthanthesumoftheoutputvoltageand inputvoltagebyasafetymargin(a10Vsafetymarginis usuallysufficient). The power dissipated by the MOSFET in a SEPIC converteris: PFET=I2SW(MAX)*RDS(ON)*DMAX +2*(VIN(MIN)+VOUT)2*IL(MAX)*CRSS*f/1A Thefirstterminthisequationrepresentstheconduction lossesinthedevice,andthesecondterm,theswitching loss.CRSSisthereversetransfercapacitance,whichis usuallyspecifiedintheMOSFETcharacteristics. For maximum efficiency, RDS(ON) and CRSS should be minimized.Fromaknownpowerdissipatedinthepower MOSFET,itsjunctiontemperaturecanbeobtainedusing thefollowingequation: TJ=TA+PFET*JA=TA+PFET*(JC+CA) TJ must not exceed the MOSFET maximum junction temperature rating. It is recommended to measure the MOSFETtemperatureinsteadystatetoensurethatabsolute maximumratingsarenotexceeded.
Thismaintainsthesameripplecurrentandenergystorage intheinductors.Thepeakinductorcurrentsare: IL1(PEAK)=IL1(MAX)+0.5*IL1 IL2(PEAK)=IL2(MAX)+0.5*IL2 TheRMSinductorcurrentsare: c2 IL1(RMS) = IL1(MAX ) * 1+ L1 12 where: cL 1 = IL1(MAX ) IL1
c 2L2 IL2(RMS) = IL 2(MAX ) * 1+ 12 where: cL 2 = IL2 (MAX ) IL2
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LT3757 applicaTions inForMaTion
SEPIC Converter: Output Diode Selection Tomaximizeefficiency,afastswitchingdiodewithalow forwarddropandlowreverseleakageisdesirable.The averageforwardcurrentinnormaloperationisequalto theoutputcurrent,andthepeakcurrentisequalto: c 1 ID(PEAK ) = 1+ * IO(MAX ) * 1- DMAX 2 CDChasnearlyarectangularcurrentwaveform.During theswitchoff-time,thecurrentthroughCDCisIIN,while approximately -IO flows during the on-time. The RMS ratingofthecouplingcapacitorisdeterminedbythefollowingequation: IRMS(CDC) > IO(MAX ) * VOUT + VD VIN(MIN)
Itisrecommendedthatthepeakrepetitivereversevoltage ratingVRRMishigherthanVOUT+VIN(MAX)byasafety margin(a10Vsafetymarginisusuallysufficient). Thepowerdissipatedbythediodeis: PD=IO(MAX)*VD andthediodejunctiontemperatureis: TJ=TA+PD*RJA TheRJAusedinthisequationnormallyincludestheRJC forthedevice,plusthethermalresistancefromtheboard, totheambienttemperatureintheenclosure.TJmustnot exceedthediodemaximumjunctiontemperaturerating. SEPIC Converter: Output and Input Capacitor Selection Theselectionsoftheoutputandinputcapacitorsofthe SEPICconverteraresimilartothoseoftheboostconverter. Please refer to the Boost Converter, Output Capacitor SelectionandBoostConverter,InputCapacitorSelection sections. SEPIC Converter: Selecting the DC Coupling Capacitor TheDCvoltageratingoftheDCcouplingcapacitor(CDC, asshowninFigure1)shouldbelargerthanthemaximum inputvoltage: VCDC>VIN(MAX)
AlowESRandESL,X5RorX7Rceramiccapacitorworks wellforCDC. INVERTING CONVERTER APPLICATIONS TheLT3757canbeconfiguredasadual-inductorinverting topology, as shown in Figure 10. The VOUT to VIN ratiois: VOUT - VD D =- VIN 1- D
incontinuousconductionmode(CCM).
L1 VIN
+
+
CDC
-
L2
CIN
-
D1 COUT
LT3757 GATE SENSE RSENSE GND M1
+
VOUT
+
3757 F09
Figure 10. A Simplified Inverting Converter
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LT3757 applicaTions inForMaTion
Inverting Converter: Switch Duty Cycle and Frequency ForaninvertingconverteroperatinginCCM,thedutycycle ofthemainswitchcanbecalculatedbasedonthenegative outputvoltage(VOUT)andtheinputvoltage(VIN). Themaximumdutycycle(DMAX)occurswhentheconverter hastheminimuminputvoltage: DMAX = VOUT - VD VOUT - VD - VIN(MIN) Afterspecifyingthemaximumoutputripple,theusercan selecttheoutputcapacitorsaccordingtothepreceding equation. TheESRcanbeminimizedbyusinghighqualityX5Ror X7Rdielectricceramiccapacitors.Inmanyapplications, ceramiccapacitorsaresufficienttolimittheoutputvoltageripple. The RMS ripple current rating of the output capacitor needstobegreaterthan: IRMS(COUT)>0.3*IL2 Inverting Converter: Selecting the DC Coupling Capacitor The DC voltage rating of the DC coupling capacitor (CDC,asshowninFigure10)shouldbelargerthanthe maximuminputvoltageminustheoutputvoltage(negativevoltage): VCDC>VIN(MAX)-VOUT CDChasnearlyarectangularcurrentwaveform.During theswitchoff-time,thecurrentthroughCDCisIIN,while approximately -IO flows during the on-time. The RMS ratingofthecouplingcapacitorisdeterminedbythefollowingequation: DMAX IRMS(CDC) > IO(MAX ) * 1- DMAX AlowESRandESL,X5RorX7Rceramiccapacitorworks wellforCDC.
Inverting Converter: Inductor, Sense Resistor, Power MOSFET, Output Diode and Input Capacitor Selections The selections of the inductor, sense resistor, power MOSFET,outputdiodeandinputcapacitorofaninverting converteraresimilartothoseoftheSEPICconverter.Please refertothecorrespondingSEPICconvertersections. Inverting Converter: Output Capacitor Selection The inverting converter requires much smaller output capacitors than those of the boost, flyback and SEPIC convertersforsimilaroutputripples.Thisisduetothefact that,intheinvertingconverter,theinductorL2isinseries withtheoutput,andtheripplecurrentflowingthroughthe outputcapacitorsarecontinuous.Theoutputripplevoltage isproducedbytheripplecurrentofL2flowingthroughthe ESRandbulkcapacitanceoftheoutputcapacitor: 1 VOUT(P - P) = IL 2 * ESRCOUT + 8 * f * COUT
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LT3757 applicaTions inForMaTion
Board Layout ThehighspeedoperationoftheLT3757demandscareful attentiontoboardlayoutandcomponentplacement.The ExposedPadofthepackageistheonlyGNDterminalof theIC,andisimportantforthermalmanagementofthe IC.Therefore,itiscrucialtoachieveagoodelectricaland thermalcontactbetweentheExposedPadandtheground planeoftheboard.FortheLT3757todeliveritsfulloutput power,itisimperativethatagoodthermalpathbeprovidedtodissipatetheheatgeneratedwithinthepackage. Itisrecommendedthatmultipleviasintheprintedcircuit boardbeusedtoconductheatawayfromtheICandinto acopperplanewithasmuchareaaspossible. To prevent radiation and high frequency resonance problems, proper layout of the components connected to the IC is essential, especially the power paths with higherdi/dt.Thefollowinghighdi/dtloopsofdifferent topologiesshouldbekeptastightaspossibletoreduce inductiveringing: * In boost configuration, the high di/dt loop contains the output capacitor, the sensing resistor, the power MOSFETandtheSchottkydiode. * In flyback configuration, the high di/dt primary loop containstheinputcapacitor,theprimarywinding,the power MOSFET and the sensing resistor. The high di/dtsecondaryloopcontainstheoutputcapacitor,the secondarywindingandtheoutputdiode. * In SEPIC configuration, the high di/dt loop contains the power MOSFET, sense resistor, output capacitor, Schottkydiodeandthecouplingcapacitor. * Ininvertingconfiguration,thehighdi/dtloopcontains powerMOSFET,senseresistor,Schottkydiodeandthe couplingcapacitor.
CIN
VIN
CC1 CC2 R1 R2 RSS RT
RC 1 2 3 4 5 10
9
R3 R4
L1
LT3757
8 7 6
CVCC
1 2 M1
8 7 6 5
RS VIAS TO GROUND PLANE
3 4
COUT2
COUT1
D1
VOUT
3757 F10
Figure 11. 8V to 16V Input, 24V/2A Output Boost Converter Suggested Layout
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LT3757 applicaTions inForMaTion
CheckthestressonthepowerMOSFETbymeasuringits drain-to-sourcevoltagedirectlyacrossthedeviceterminals (referencethegroundofasinglescopeprobedirectlyto the source pad on the PC board). Beware of inductive ringing,whichcanexceedthemaximumspecifiedvoltage ratingoftheMOSFET.Ifthisringingcannotbeavoided, and exceeds the maximum rating of the device, either chooseahighervoltagedeviceorspecifyanavalancheratedpowerMOSFET. Thesmall-signalcomponentsshouldbeplacedawayfrom highfrequencyswitchingnodes.Foroptimumloadregulationandtrueremotesensing,thetopoftheoutputvoltage sensingresistordividershouldconnectindependentlyto thetopoftheoutputcapacitor(Kelvinconnection),staying awayfromanyhighdV/dttraces.PlacethedividerresistorsneartheLT3757inordertokeepthehighimpedance FBXnodeshort. Figure11showsthesuggestedlayoutofthe8Vto16V Input,24V/2AOutputBoostConverter. Recommended Component Manufacturers Some of the recommended component manufacturers arelistedinTable2.
Table 2. Recommended Component Manufacturers
VENDOR AVX BHElectronics Coilcraft CooperBussmann Diodes,Inc Fairchild General Semiconductor InternationalRectifier IRC Kemet MagneticsInc Microsemi Murata-Erie Nichicon OnSemiconductor Panasonic Sanyo Sumida TaiyoYuden TDK Thermalloy Tokin Toko UnitedChemicon Vishay/Dale Vishay/Siliconix Vishay/Sprague WurthElectronik Zetex COMPONENTS Capacitors Inductors, Transformers Inductors Inductors Diodes MOSFETs Diodes MOSFETs,Diodes SenseResistors Capacitors ToroidCores Diodes Inductors, Capacitors Capacitors Diodes Capacitors Capacitors Inductors Capacitors Capacitors, Inductors HeatSinks Capacitors Inductors Capacitors Resistors MOSFETs Capacitors Inductors Small-Signal Discretes WEB ADDRESS avx.com bhelectronics.com coilcraft.com bussmann.com diodes.com fairchildsemi.com generalsemiconductor.com irf.com irctt.com kemet.com mag-inc.com microsemi.com murata.co.jp nichicon.com onsemi.com panasonic.com sanyo.co.jp sumida.com t-yuden.com component.tdk.com aavidthermalloy.com nec-tokinamerica.com tokoam.com chemi-com.com vishay.com vishay.com vishay.com we-online.com zetex.com
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LT3757 Typical applicaTions
3.3V Input, 5V/10A Output Boost Converter
L1 0.5H CIN 22F 6.3V 2 VIN INTVCC SHDN/UVLO 34k SYNC VIN 3.3V
49.9k
LT3757
GATE FBX RT SS VC SENSE GND
CVCC 4.7F 10V X5R
D1 M1
VOUT 5V 10A 34k 1%
22
+
COUT1 150F 6.3V 4 COUT2 22F 6.3V X5R 4
41.2k 300kHz 0.1F
6.8k 22nF 2.2nF
0.004 1W 15.8k 1%
CIN: TAIYO YUDEN JMK325BJ226MM COUT1: PANASONIC EEFUEOJ151R COUT2: TAIYO YUDEN JMK325BJ226MM
D1: MBRB2515L L1: VISHAY SILICONIX IHLP-5050FD-01 M1: VISHAY SILICONIX SI4448DY
3757 TA02a
Efficiency vs Output Current
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 0.001 0.01 0.1 1 10
3757 TA02b
OUTPUT CURRENT (A)
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LT3757 Typical applicaTions
8V to 16V Input, 24V/2A Output Boost Converter
VIN 8V TO 16V CIN 10F 25V X5R
R3 200k R4 43.2k SYNC
VIN SHDN/UVLO
L1 10H D1 GATE SENSE M1 R2 226k 1% VOUT 24V 2A
LT3757
RT SS VC RT 41.2k 300kHz CC2 100pF CSS 0.1F RC 22k CC1 6.8nF GND
FBX INTVCC RS 0.01 1W
+
COUT1 47F 35V 4
CVCC 4.7F 10V X5R
R1 16.2k 1%
COUT2 10F 25V X5R
CIN, COUT2: MURATA GRM31CR61E106KA12 COUT1: KEMET T495X476K035AS D1: ON SEMI MBRS340T3G L1: VISHAY SILICONIX IHLP-5050FD-01 10H M1: VISHAY SILICONIX Si4840BDP
3757 TA03a
Efficiency vs Output Current
100 90 80 EFFICIENCY (%) 70 60 50 40 30 0.001 0.1 1 0.01 OUTPUT CURRENT (A) 10
3757 TA03b
Load Step Response at VIN = 12V
VIN = 8V VIN = 16V
VOUT 500mV/DIV (AC)
IOUT 1.6A 1A/DIV 0.4A 500s/DIV
3757 TA03c
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LT3757 Typical applicaTions
High Voltage Flyback Power Supply
DANGER! HIGH VOLTAGE OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY VIN 5V TO 12V T1 1:10 CIN 150F 6.3V 2 105k SHDN/UVLO 46.4k SYNC VIN INTVCC CVCC 47F 25V X5R 22 22 220pF VSW M1 D1 VOUT 350V 10mA 1.50M 1% 1M 1% 1M 1%
* *
LT3757
GATE
COUT 68nF 2
RT SS VC 140k 100kHz 0.1F 100pF
GND
SENSE FBX
10nF 6.8k 22nF 0.02
16.2k 1%
CIN : MURATA GRM32DR61C106K COUT : TDK C3225X7R2J683K D1: VISHAY SILICONIX GSD2004S DUAL DIODE CONNECTED IN SERIES M1: VISHAY SILICONIX Si7850DP T1: TDK DCT15EFD-U44S003
3757 TA04a
Start-Up Waveforms
VOUT 5V/DIV (AC)
Switching Waveforms
VSW 20V/DIV VOUT 100V/DIV
3757 TA04b 3757 TA04c
2ms/DIV
5s/DIV
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0
LT3757 Typical applicaTions
5.5V to 36V Input, 12V/2A Output SEPIC Converter
VIN 5.5V TO 36V CIN 4.7F 50V 2 105k VIN SHDN/UVLO 46.4k SYNC L1A
*
IL1A
CDC 4.7F 50V, X5R, 2
D1
LT3757
VSW GATE SENSE M1 IL1B L1B
VOUT 12V 2A 105k 1%
*
RT SS VC 41.2k 300kHz 0.1F
FBX GND INTVCC
0.008 1W
+
COUT1 47F 20V 2 COUT2 10F 25V X5R
3757 TA05a
10k 6.8nF
4.7F 10V X5R
15.8k 1%
CIN, CDC: TAIYO YUDEN UMK316BJ475KL COUT1: KEMET T495X476K020AS COUT2: TAIYO YUDEN TMK432BJ106MM D1: ON SEMI MBRS360T3G L1A, L1B: COILTRONICS DRQ127-3R3 (*COUPLED INDUCTORS) M1: VISHAY SILICONIX Si7460DP
Efficiency vs Output Current
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 0.001 0.1 1 0.01 OUTPUT CURRENT (A) 10
3757 TA05b
Load Step Waveforms
VIN = 8V VIN = 16V
VOUT 200mV/DIV (AC)
IOUT 1.6A 1A/DIV 0.4A 500s/DIV
3757 TA05c
Start-Up Waveforms
VIN = 12V
Frequency Foldback Waveforms When Output Short-Circuits
VOUT 10V/DIV VSW 20V/DIV VIN = 12V
VOUT 5V/DIV
IL1A + IL1B 5A/DIV 2ms/DIV
3757 TA05d
IL1A + IL1B 5A/DIV
3757 TA05e
50s/DIV
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LT3757 Typical applicaTions
5V to 12V Input, 12V/0.2A Output SEPIC Converter
VIN 5V TO 12V + CIN2 47F 16V CIN1 1F 16V, X5R
105k
VIN SHDN/UVLO
*
T1 1,2,3,4
CDC1 4.7F 16V, X5R
D1 CDC2 4.7F 16V X5R 1.05k 1% 158 1% COUT2 4.7F 16V, X5R 3 COUT2 4.7F 16V, X5R 3
3757 TA06
46.4k SYNC
LT3757
GATE SENSE
M1
VOUT1 12V 0.4A
5 * D2 6 *
RT SS VC 30.9k 400kHz 0.1F 100pF
FBX GND INTVCC
0.02
GND
22k 6.8nF
VOUT2 -12V 0.4A
CVCC 4.7F 10V X5R
D1, D2: MBRS140T3 T1: COILTRONICS VP1-0076 (*PRIMARY = 4 WINDINGS IN PARALLEL) M1: SILICONIX/VISHAY Si4840BDY
Nonisolated Inverting SLIC Supply
VP5-0155 (PRIMARY = 3 WINDINGS IN PARALLEL) D1 DFLS160 CIN 22F 25V, X5R 2 GND R2 105k R1 46.4k SYNC VIN SHDN/UVLO LT3757 C2 10F 50V X5R T1 1,2,3 * * D2 DFLS160 C4 22F 25V X5R 4 C3 22F 25V X5R
VIN 5V TO 16V
GATE SENSE
M1 Si7850DP
RT SS VC 63.4k 200kHz 0.1F 100pF
FBX GND INTVCC 0.012 0.5W *
5
COUT 3.3F 100V
VOUT1 -24V 200mA
9.1k 10nF
D3 DFLS160 C5 22F 25V X5R
CVCC 4.7F 10V, X5R
6 *
15.8k
464k
3757 TA07
VOUT1 -72V 200mA
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LT3757 package DescripTion
(ReferenceLTCDWG#05-08-1699RevB)
DD Package 10-Lead Plastic DFN (3mm x 3mm)
0.70 0.05
3.55 0.05 1.65 0.05 2.15 0.05 (2 SIDES) PACKAGE OUTLINE 0.25 0.05
0.50 BSC 2.38 0.05 (2 SIDES) R = 0.125 TYP 6 0.40 10 0.10
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
3.00 0.10 (4 SIDES) PIN 1 TOP MARK (SEE NOTE 6)
1.65 0.10 (2 SIDES)
5 0.200 REF 0.75 0.05 2.38 0.10 (2 SIDES)
1
(DD) DFN REV B 0309
0.25 0.05 0.50 BSC
0.00 - 0.05
BOTTOM VIEW--EXPOSED PAD
NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
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LT3757 package DescripTion
MSE Package 10-Lead Plastic MSOP Exposed Die Pad ,
(ReferenceLTCDWG#05-08-1664RevC)
BOTTOM VIEW OF EXPOSED PAD OPTION
2.794 (.110
0.102 .004)
0.889 (.035
0.127 .005)
1
2.06 0.102 (.081 .004) 1.83 0.102 (.072 .004) 0.05 REF
0.29 REF
5.23 (.206) MIN
2.083 (.082
0.102 3.20 - 3.45 .004) (.126 - .136)
10
DETAIL "B" CORNER TAIL IS PART OF DETAIL "B" THE LEADFRAME FEATURE. FOR REFERENCE ONLY NO MEASUREMENT PURPOSE 0.497 0.076 (.0196 .003) REF
0.50 0.305 0.038 (.0197) (.0120 .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT
3.00 0.102 (.118 .004) (NOTE 3)
10 9 8 7 6
4.90 0.152 (.193 .006) 0.254 (.010)
GAUGE PLANE DETAIL "A" 0 - 6 TYP
3.00 0.102 (.118 .004) (NOTE 4)
12345 0.53 0.152 (.021 .006)
DETAIL "A"
1.10 (.043) MAX
0.86 (.034) REF
0.18 (.007)
SEATING PLANE
0.50 (.0197) NOTE: BSC 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.17 - 0.27 (.007 - .011) TYP
0.1016 (.004
0.0508 .002)
MSOP (MSE) 0908 REV C
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LT3757 revision hisTory
REV B DATE 3/10 DESCRIPTION DeletedBulletfromFeaturesandLastLineofDescription UpdatedEntirePagetoAddH-GradeandMilitaryGrade UpdatedElectricalCharacteristicsNotesandTypicalPerformanceCharacteristicsforH-GradeandMilitaryGrade RevisedTA04aandReplacedTA04cinTypicalApplications UpdatedRelatedParts
(Revision history begins at Rev B)
PAGE NUMBER 1 2 4to6 30 36
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However,noresponsibilityisassumedforitsuse.LinearTechnologyCorporationmakesnorepresentationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
LT3757 Typical applicaTion
High Efficiency Inverting Power Supply
VIN 5V TO 15V CIN 47F 16V X5R R2 105k R1 46.4k SYNC VIN SHDN/UVLO
Efficiency vs Output Current
100 90 L2 VOUT -5V 3A to 5A 84.5k 80 EFFICIENCY (%) 70 60 50 40 30 16k COUT 100F 6.3V, X5R 2
3757 TA08a
L1
*
CDC 47F 25V, X5R
LT3757
VIN = 5V VIN = 16V
GATE SENSE
M1 Si7848BDP 0.006 1W
RT SS VC 41.2k 300kHz 0.1F 9.1k 10nF GND
D1 MBRD835L
FBX INTVCC CVCC 4.7F 10V X5R
*
20 10 0.001 0.1 1 0.01 OUTPUT CURRENT (A) 10
3757 TA08b
L1, L2: COILTRONICS DRQ127-3R3 (*COUPLED INDUCTORS)
relaTeD parTs
PART NUMBER LT3758 DESCRIPTION Boost,Flyback,SEPICandInvertingController COMMENTS 2.9VVIN100V,CurrentModeControl,100kHzto1MHzProgrammable OperationFrequency,3mmx3mm10-LeadDFNand10-LeadMSOP-E Packages 3VVIN40V,NoOpto-IsolatororThirdWindingRequired,Upto7W, 16-LeadMSOP-EPackage AdjustableSwitchingFrequency,2.5VVIN36V,BurstMode(R)Operationat LightLoads 2.75VVIN9.8V,23-LeadThinSotTMand2mmx3mm8-LeadDFN Packages IdealforVINfrom4.5Vto36VLimitedbyExternalComponents,Upto60W, CurrentModeControl VIN16Vto75VLimitedbyExternalComponents,Upto60W,CurrentMode Control VINandVOUTLimitedOnlybyExternalComponents,6-LeadThinSotPackage VINandVOUTLimitedOnlybyExternalComponents,3mmx3mm10-Lead DFN,10-LeadMSOP-EPackages
LT3573 LTC1871/LTC1871-1/ LTC1871-7 LTC3872 LT3837 LT3825 LTC3803/LTC3803-3/ LTC3803-5 LTC3805/LTC3805-5
IsolatedFlybackSwitchingRegulatorwith60V IntegratedSwitch Boost,FlybackandSEPICController,NoRSENSETM, LowQuiescentCurrent Boost,Flyback,SEPICController IsolatedNo-OptoSynchronousFlybackController IsolatedNo-OptoSynchronousFlybackController 200kHzFlybackDC/DCController AdjustableFixed70kHzto700kHzOperating FrequencyFlybackController
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Linear Technology Corporation
(408)432-1900 FAX:(408)434-0507 www.linear.com
LT 0310 REV B * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
LINEAR TECHNOLOGY CORPORATION 2008

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