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LTC1627 Monolithic Synchronous Step-Down Switching Regulator
FEATURES
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DESCRIPTION
The LTC(R)1627 is a monolithic current mode synchronous buck regulator using a fixed frequency architecture. The operating supply range is from 8.5V down to 2.65V, making it suitable for one or two lithium-ion battery-powered applications. Burst Mode operation provides high efficiency at low load currents. 100% duty cycle provides low dropout operation, which extends operating time in battery-operated systems. The operating frequency is internally set at 350kHz, allowing the use of small surface mount inductors. For switching noise sensitive applications it can be externally synchronized up to 525kHz. The SYNC/FCB control pin guarantees regulation of secondary windings regardless of load on the main output by forcing continuous operation. Burst Mode operation is inhibited during synchronization or when the SYNC/FCB pin is pulled low to reduce noise and RF interference. Soft-start is provided by an external capacitor. Optional bootstrapping enhances the internal switch drive for single lithium-ion cell applications. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode, saving components and board space. The LTC1627 comes in an 8-lead SO package.
, LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation.
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High Efficiency: Up to 96% Constant Frequency 350kHz Operation 2.65V to 8.5V VIN Range VOUT from 0.8V to VIN, IOUT to 500mA No Schottky Diode Required Synchronizable Up to 525kHz Selectable Burst ModeTM Operation Low Dropout Operation: 100% Duty Cycle Precision 2.5V Undervoltage Lockout Secondary Winding Regulation Current Mode Operation for Excellent Line and Load Transient Response Low Quiescent Current: 200A Shutdown Mode Draws Only 15A Supply Current 1.5% Reference Accuracy Available in 8-Lead SO Package
APPLICATIONS
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Cellular Telephones Portable Instruments Wireless Modems RF Communications Distributed Power Systems Scanners Single and Dual Cell Lithium
TYPICAL APPLICATION
1 47pF CSS 0.1F 2 3 4 ITH SYNC/FCB 8 7 6 5 L1 15H COUT + 100F 6.3V VOUT 3.3V 249k VIN 2.8V* TO 8.5V CIN 22F 16V
EFFICIENCY (%)
100 VOUT = 3.3V 95 VIN = 6V 90 85 80 75 70 VIN = 8.4V VIN = 3.6V
RUN/SS VDR LTC1627 VFB GND VIN SW
+
*VOUT CONNECTED TO VIN FOR 2.8V < VIN < 3.3V
80.6k
1627 F01a
1
Figure 1a. High Efficiency Step-Down Converter
Figure 1b. Efficiency vs Output Load Current
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10 100 OUTPUT CURRENT (mA)
1000
1627 F01b
1
LTC1627
ABSOLUTE MAXIMUM RATINGS
(Note 1)
PACKAGE/ORDER INFORMATION
TOP VIEW ITH 1 RUN/SS 2 VFB 3 GND 4 8 7 6 5 SYNC/FCB VDR VIN SW
Input Supply Voltage ................................ - 0.3V to 10V Driver Supply Voltage (VIN - VDR) ........... - 0.3V to 10V ITH Voltage .................................................. - 0.3V to 5V Run/SS, VFB Voltages ................................ - 0.3V to VIN Sync/FCB Voltage ...................................... - 0.3V to VIN VDR Voltage (VIN 5V) ............................... - 5V to 0.3V P-Channel Switch Source Current (DC) .............. 800mA N-Channel Switch Sink Current (DC) .................. 800mA Peak SW Sink and Source Current.......................... 1.5A Operating Ambient Temperature Range Commercial ............................................ 0C to 70C Industrial ........................................... - 40C to 85C Junction Temperature (Note 2) ............................. 125C Storage Temperature Range ................. - 65C to 150C Lead Temperature (Soldering, 10 sec).................. 300C
ORDER PART NUMBER LTC1627CS8 LTC1627IS8 S8 PART MARKING 1627 1627I
S8 PACKAGE 8-LEAD PLASTIC SO TJMAX = 125C, JA = 110C/ W
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
SYMBOL IVFB VFB VOVL VFB VLOADREG IS PARAMETER Feedback Current Regulated Feedback Voltage Output Overvoltage Lockout Reference Voltage Line Regulation Output Voltage Load Regulation Input DC Bias Current Synchronized Burst Mode Operation Shutdown Shutdown Run/SS Threshold Soft-Start Current Source Auxiliary Feedback Threshold Auxiliary Feedback Current Oscillator Frequency Undervoltage Lockout RDS(ON) of P-Channel FET RDS(ON) of N-Channel FET SW Leakage VRUN/SS = 0V CONDITIONS (Note 3) (Note 3)
TA = 25C, VIN = 5V unless otherwise specified.
MIN
q
TYP 20 0.80 60 0.002 0.5 - 0.5 450 200 15 6
MAX 60 0.812 110 0.01 0.8 - 0.8
UNITS nA V mV %/V % % A A A A V A V A kHz kHz V V nA
0.788 20
VOVL = VOVL - VFB VIN = 2.8V to 8.5V (Note 3) ITH Sinking 2A (Note 3) ITH Sourcing 2A (Note 3) (Note 4) VIN = 8.5V, VOUT = 3.3V, Frequency = 525kHz VITH = 0V, VIN = 8.5V, VSYNC/FCB = Open VRUN/SS = 0V, 2.65V < VIN < 8.5V VRUN/SS = 0V, VIN < 2.65V
320 35 1.0 3.3 0.835 2.5 385 2.65 2.80 0.7 0.8 1000
VRUN/SS IRUN/SS VSYNC/FCB ISYNC/FCB fOSC VUVLO RPFET RNFET ILSW
0.4 1.2 0.755 0.5 315
q q
0.7 2.25 0.8 1.5 350 35 2.50 2.65 0.5 0.6 10
VSYNC/FCB Ramping Negative VSYNC/FCB = 0V VFB = 0.8V VFB = 0V VIN Ramping Down from 3V VIN Ramping Up from 0V (VIN - VDR) = 5V, ISW = 100mA ISW = - 100mA VRUN/SS = 0V
2.4
The q denotes specifications which apply over the full operating temperature range. Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD * 110C/W)
Note 3: The LTC1627 is tested in a feedback loop that servos VFB to the balance point for the error amplifier (VITH = 0.8V). Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency.
2
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WW
W
LTC1627 TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Input Voltage
100
100 95
95
ILOAD = 100mA ILOAD = 300mA ILOAD = 10mA
EFFICIENCY (%)
EFFICIENCY (%)
90
VDR = - VIN 85 80 75 70 VIN = 3.6V VOUT = 2.5V L = 15H Burst Mode OPERATION 1 10 100 OUTPUT CURRENT (mA) 1000
1627 G02
EFFICIENCY (%)
85
80
VOUT = 2.5V L = 15H VDR = 0V Burst Mode OPERATION 0 2 6 4 INPUT VOLTAGE (V) 8 10
1627 G01
75
Efficiency vs Load Current
100 VIN = 2.8V 95 EFFICIENCY (%) 90 85 VIN = 7.2V 80 75 70 1 10 100 OUTPUT CURRENT (mA) 1000
1627 G04
UNDERVOLTAGE LOCKOUT THRESHOLD (V)
BATTERY VOLTAGE (V)
VIN = 3.6V
VOUT = 2.5V L = 15H VDR = 0V Burst Mode OPERATION
Supply Current in Shutdown vs Input Voltage
22
SUPPLY CURRENT IN SHUTDOWN (A)
VRUN/SS = 0V TJ = 85C TJ = 25C
0.808 0.806
FORCED CONTINUOUS THRESHOLD VOLTAGE (V)
20
16 14 12 10 8 6 4 2.5 3.5
REFERENCE VOLTAGE (V)
18
TJ = - 40C
4.5 5.5 6.5 INPUT VOLTAGE (V)
UW
7.5
1627 G07
Efficiency vs Load Current
100 95 VDR = 0V 90 90 85
Efficiency vs Load Current
VIN = 3.6V VOUT = 2.5V L = 15H VDR = 0V Burst Mode OPERATION
SYNCHRONIZED AT 525kHz 80 FORCED CONTINUOUS 75 70 1 10 100 OUTPUT CURRENT (mA) 1000
1627 G03
Undervoltage Lockout Threshold vs Temperature
2.75 2.70 2.65 2.60 2.55 2.50 2.45 2.40 2.35 2.30 - 50 - 25 0 25 50 75 TEMPERATURE (C) 100 125 VIN RAMPING DOWN VIN RAMPING UP 550 500 450 400 350 300 250 200
DC Supply Current* vs Input Voltage
TJ = 25C VOUT = 1.8V SYNCHRONIZED AT 525kHz
Burst Mode OPERATION 150 100 2.5 3.5 4.5 5.5 6.5 INPUT VOLTAGE (V) 7.5 8.5
1627 G05
1627 G06
*DOES NOT INCLUDE GATE CHARGE CURRENT
Reference Voltage vs Temperature
VIN = 5V
Forced Continuous Threshold Voltage vs Temperature
0.808 0.806 0.804 0.802 0.800 0.798 0.796 0.794 0.792 0.790 - 50 - 25 0 25 50 75 TEMPERATURE (C) 100 125 VIN = 5V
0.804 0.802 0.800 0.798 0.796 0.794 0.792
8.5
0.790 - 50
- 25
0 25 50 75 TEMPERATURE (C)
100
125
1627 G08
1627 G09
3
LTC1627 TYPICAL PERFOR A CE CHARACTERISTICS
Oscillator Frequency vs Temperature
390 380 VIN = 5V VSYNC/FCB = 0V 390 380
MAXIMUM OUTPUT LOAD CURRENT (mA)
OSCILLATOR FREQUENCY (kHz)
OSCILLATOR FREQUENCY (kHz)
370 360 350 340 330 320 310 300 - 50 - 25 0 25 50 75 TEMPERATURE (C) 100 125
Switch Leakage Current vs Temperature
1800
SYNCHRONOUS SWITCH LEAKAGE (nA)
1600 1400 1200 1000 800 600 400 200
VIN = 8.4V VDR = 0V
SWITCH RESISTANCE ()
SWITCH RESISTANCE ()
SYNCHRONOUS SWITCH
0 - 50
- 25
0 25 50 75 TEMPERATURE (C)
Load Step Transient Response
ITH 0.5V/DIV ITH 0.5V/DIV
VOUT 50mV/DIV AC COUPLED ILOAD 500mA/DIV 25s/DIV VIN = 5V VOUT = 3.3V L = 15H CIN = 22F COUT = 100F ILOAD = 0mA TO 500mA Burst Mode OPERATION
1627 G16
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UW
1627 G10
Oscillator Frequency vs Input Voltage
VSYNC/FCB = 0V
Maximum Output Load Current vs Input Voltage
1100 1000 900 800 700 600 500 400 300 200 2.5 3.5 4.5 5.5 6.5 INPUT VOLTAGE (V) VOUT = 2.5V L = 15H 7.5 8.5 VDR = 0V VDR = -VIN
370 360 350 340 330 320 310 300 2.5 3.5 4.5 5.5 6.5 INPUT VOLTAGE (V) 7.5 8.5
1627 G11
1627 G12
Switch Resistance vs Temperature
0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 - 50 - 25 0 25 50 75 TEMPERATURE (C) 100 125 MAIN SWITCH VIN = 5V VDR = 0V SYNCHRONOUS SWITCH 0.9 0.8 0.7
Switch Resistance vs Input Voltage
VDR = 0V
SYNCHRONOUS SWITCH 0.6 0.5 MAIN SWITCH 0.4 0.3 0.2 0.1 0 2.5 3.5 4.5 5.5 6.5 INPUT VOLTAGE (V) 7.5 8.5
MAIN SWITCH 100 125
1627 G13
1627 G14
1627 G15
Load Step Transient Response
SW 5V/DIV VOUT 20mV/DIV AC COUPLED ILOAD 200mA/DIV 25s/DIV VIN = 5V VOUT = 3.3V L = 15H CIN = 22F COUT = 100F ILOAD = 0mA TO 500mA FORCED CONTINUOUS MODE
1627 G17
Burst Mode Operation
VOUT 50mV/DIV AC COUPLED ILOAD 500mA/DIV
10s/DIV VIN = 5V VOUT = 3.3V L = 15H CIN = 22F COUT = 100F ILOAD = 50mA
1627 G18
LTC1627
PIN FUNCTIONS
ITH (Pin 1): Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0V to 1.2V. RUN/SS (Pin 2): Combination of Soft-Start and Run Control Inputs. A capacitor to ground at this pin sets the ramp time to full current output. The time is approximately 0.5s/F. Forcing this pin below 0.4V shuts down all the circuitry. VFB (Pin 3): Feedback Pin. Receives the feedback voltage from an external resistive divider across the output. GND (Pin 4): Ground Pin. SW (Pin 5): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. VIN (Pin 6): Main Supply Pin. Must be closely decoupled to GND, Pin 4. VDR (Pin 7): Top Driver Return Pin. This pin can be bootstrapped to go below ground to improve efficiency at low VIN (see Applications Information). SYNC/FCB (Pin 8): Multifunction Pin. This pin performs three functions: 1) secondary winding feedback input, 2) external clock synchronization and 3) Burst Mode operation or forced continuous mode select. For secondary winding applications connect a resistive divider from the secondary output. To synchronize with an external clock apply a TTL/CMOS compatible clock with a frequency between 385kHz and 525kHz. To select Burst Mode operation, float the pin or tie it to VIN. Grounding Pin 8 forces continuous operation (see Applications Information).
FUNCTIONAL DIAGRA
BURST DEFEAT X 1.5A SYNC/FCB 8 Y
Y = "0" ONLY WHEN X IS A CONSTANT "1"
VIN
SLOPE COMP OSC 0.4V
0.6V VFB 3
VIN 0.8V
+
EA 0.12V ITH 1 S RUN/SOFT START R Q Q
-
0.8V REF 2.25A VIN RUN/SS 2 UVLO TRIP = 2.5V
+
OVDET 0.86V SHUTDOWN 0.8V
- +
IRCMP 5 SW 4 GND
1627 BD
FCB
+
-
+
FREQ SHIFT
-
W
+
-
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+ -
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VIN
VIN
6 VIN EN SLEEP
-
+
6
ICOMP
BURST
SWITCHING LOGIC AND BLANKING CIRCUIT
ANTISHOOT-THRU
7 VDR
-
5
LTC1627
OPERATIO
Main Control Loop The LTC1627 uses a constant frequency, current mode step-down architecture. Both the main and synchronous switches, consisting of top P-channel and bottom N-channel power MOSFETs, are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch is controlled by the voltage on the ITH pin, which is the output of error amplifier EA. The VFB pin, described in the Pin Functions section, allows EA to receive an output feedback voltage from an external resistive divider. When the load current increases, it causes a slight decrease in the feedback voltage relative to the 0.8V reference, which, in turn, causes the ITH voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse as indicated by the current reversal comparator IRCMP, or the beginning of the next cycle. The main control loop is shut down by pulling the RUN/SS pin low. Releasing RUN/SS allows an internal 2.25A current source to charge soft-start capacitor CSS. When CSS reaches 0.7V, the main control loop is enabled with the ITH voltage clamped at approximately 5% of its maximum value. As CSS continues to charge, ITH is gradually released, allowing normal operation to resume. Comparator OVDET guards against transient overshoots > 7.5% by turning the main switch off and turning the synchronous switch on. With the synchronous switch turned on, the output is crowbarred. This may cause a large amount of current to flow from VIN if the main switch is damaged, blowing the system fuse. Burst Mode Operation The LTC1627 is capable of Burst Mode operation in which the internal power MOSFETs operate intermittently based on load demand. To enable Burst Mode operation, simply allow the SYNC/FCB pin to float or connect it to a logic high. To disable Burst Mode operation and enable forced continuous mode, connect the SYNC/FCB pin to GND. In this mode, the efficiency is lowest at light loads, but becomes comparable to Burst Mode operation when the
6
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(Refer to Functional Diagram)
output load exceeds 100mA. The threshold voltage between Burst Mode operation and forced continuous mode is 0.8V. This can be used to assist in secondary winding regulation as described in Auxiliary Winding Control Using SYNC/FCB Pin in the Applications Information section. When the converter is in Burst Mode operation, the peak current of the inductor is set to approximately 200mA, even though the voltage at the ITH pin indicates a lower value. The voltage at the ITH pin drops when the inductor's average current is greater than the load requirement. As the ITH voltage drops below 0.12V, the BURST comparator trips, causing the internal sleep line to go high and turn off both power MOSFETs. In sleep mode, both power MOSFETs are held off and the internal circuitry is partially turned off, reducing the quiescent current to 200A. The load current is now being supplied from the output capacitor. When the output voltage drops, causing ITH to rise above 0.22V, the top MOSFET is again turned on and this process repeats. Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator is reduced to about 35kHz, 1/10 the nominal frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing runaway. The oscillator's frequency will progressively increase to 350kHz (or the synchronized frequency) when VFB rises above 0.3V. Frequency Synchronization The LTC1627 can be synchronized with an external TTL/CMOS compatible clock signal. The frequency range of this signal must be from 385kHz to 525kHz. Do not attempt to synchronize the LTC1627 below 385kHz as this may cause abnormal operation and an undesired frequency spectrum. The top MOSFET turn-on follows the rising edge of the external source. When the LTC1627 is clocked by an external source, Burst Mode operation is disabled; the LTC1627 then operates in PWM pulse skipping mode. In this mode, when the output load is very low, current comparator ICOMP remains tripped for more than one cycle and forces the main switch to stay off for the same number of cycles. Increasing the output
LTC1627
OPERATIO
load slightly allows constant frequency PWM operation to resume. Frequency synchronization is inhibited when the feedback voltage VFB is below 0.6V. This prevents the external clock from interfering with the frequency foldback for shortcircuit protection. Dropout Operation When the input supply voltage decreases toward the output voltage, the duty cycle increases toward the maximum on-time. Further reduction of the supply voltage forces the main switch to remain on for more than one cycle until it reaches 100% duty cycle. The output voltage will then be determined by the input voltage minus the voltage drop across the P-channel MOSFET and the inductor. In Burst Mode operation or pulse skipping mode operation (externally synchronized) with the output lightly loaded, the LTC1627 transitions through continuous mode as it enters dropout. Undervoltage Lockout A precision undervoltage lockout shuts down the LTC1627 when VIN drops below 2.5V, making it ideal for single lithium-ion battery applications. In lockout, the LTC1627 draws only several microamperes, which is low enough to prevent deep discharge and possible damage to the lithiumion battery nearing its end of charge. A 150mV hysteresis ensures reliable operation with noisy supplies. Low Supply Operation The LTC1627 is designed to operate down to 2.65V supply voltage. At this voltage the converter is most likely to be running at high duty cycles or in dropout where the main switch is on continuously. Hence, the I2R loss is due mainly to the RDS(ON) of the P-channel MOSFET. See Efficiency Considerations in the Applications Information section. When VIN is low (< 4.5V) the RDS(ON) of the P-channel MOSFET can be lowered by driving its gate below ground. The top P-channel MOSFET driver makes use of a floating return pin, VDR, to allow biasing below GND. A simple charge pump bootstrapped to the SW pin realizes a negative voltage at the VDR pin as shown in Figure 2. Using
MAXIMUM INDUCTOR PEAK CURRENT (mA)
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VDR LTC1627 VIN D1 L1 SW C2 0.1F D2 C1 0.1F VIN < 4.5V VOUT COUT 100F
1627 F02
+
Figure 2. Using a Charge Pump to Bias VDR
the charge pump at VIN 4.5V is not recommended to ensure that (VIN - VDR) does not exceed its absolute maximum voltage. When VIN decreases to a voltage close to VOUT, the loop may enter dropout and attempt to turn on the P-channel MOSFET continuously. When the VDR charge pump is enabled, a dropout detector counts the number of oscillator cycles that the P-channel MOSFET remains on, and periodically forces a brief off period to allow C1 to recharge. 100% duty cycle is allowed when VDR is grounded. Slope Compensation and Inductor Peak Current Slope compensation provides stability by preventing subharmonic oscillations. It works by internally adding a ramp to the inductor current signal at duty cycles in excess of 40%. As a result, the maximum inductor peak current is lower for VOUT/VIN > 0.4 than when VOUT/VIN < 0.4. See the inductor peak current as a function of duty cycle graph in Figure 3. The worst-case peak current reduction occurs with the oscillator synchronized at its minimum frequency, i.e., to a clock just above the oscillator free-running
950 900 850 800 750 700 650 600 550 500 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%)
1627 F03
WITHOUT EXTERNAL CLOCK SYNC
WORST CASE EXTERNAL CLOCK SYNC
VIN = 5V
Figure 3. Maximum Inductor Peak Current vs Duty Cycle
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LTC1627
APPLICATIONS INFORMATION
frequency. The actual reduction in average current is less than for peak current. The basic LTC1627 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Inductor Value Calculation The inductor selection will depend on the operating frequency of the LTC1627. The internal preset frequency is 350kHz, but can be externally synchronized up to 525kHz. The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. However, operating at a higher frequency generally results in lower efficiency because of internal gate charge losses. The inductor value has a direct effect on ripple current. The ripple current IL decreases with higher inductance or frequency and increases with higher VIN or VOUT. size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Kool M (from Magnetics, Inc.) is a very good, low loss core material for toroids with a "soft" saturation characteristic. Molypermalloy is slightly more efficient at high (>200kHz) switching frequencies but quite a bit more expensive. Toroids are very space efficient, especially when you can use several layers of wire, while inductors wound on bobbins are generally easier to surface mount. New designs for surface mount are available from Coiltronics, Coilcraft and Sumida. CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN required IRMS IMAX
IL =
( )( )
V VOUT 1 - OUT VIN fL 1
Accepting larger values of IL allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is IL = 0.4(IMAX). The inductor value also has an effect on Burst Mode operation. The transition to low current operation begins when the inductor current peaks fall to approximately 200mA. Lower inductor values (higher IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy, or Kool M(R) cores. Actual core loss is independent of core
Kool M is a registered trademark of Magnetics, Inc.
8
U
W
U
U
(1)
[
VOUT VIN - VOUT VIN
(
)]
1/ 2
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT /2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question.
LTC1627
APPLICATIONS INFORMATION
The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple VOUT is determined by:
0.8V VOUT 8.5V R2 VFB LTC1627 GND
1627 F04
1 VOUT IL ESR + 4fCOUT
where f = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. The output ripple is highest at maximum input voltage since IL increases with input voltage. For the LTC1627, the general rule for proper operation is: COUT required ESR < 0.25 Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR/size ratio of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo POSCAP, KEMET T510 and T495 series, Nichicon PL series and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Output Voltage Programming The output voltage is set by a resistive divider according to the following formula:
R2 VOUT = 0.8V 1 + R1
The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 4.
U
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R1
Figure 4. Setting the LTC1627 Output Voltage
Run/Soft-Start Function The RUN/SS pin is a dual purpose pin that provides the soft-start function and a means to shut down the LTC1627. Soft-start reduces surge currents from VIN by gradually increasing the internal current limit. Power supply sequencing can also be accomplished using this pin. An internal 2.25A current source charges up an external capacitor CSS. When the voltage on RUN/SS reaches 0.7V the LTC1627 begins operating. As the voltage on RUN/SS continues to ramp from 0.7V to 1.8V, the internal current limit is also ramped at a proportional linear rate. The current limit begins at 25mA (at VRUN/SS 0.7V) and ends at the Figure 3 value (VRUN/SS 1.8V). The output current thus ramps up slowly, charging the output capacitor. If RUN/SS has been pulled all the way to ground, there will be a delay before the current starts increasing and is given by:
tDELAY =
0.7CSS 2.25A
Pulling the RUN/SS pin below 0.4V puts the LTC1627 into a low quiescent current shutdown (IQ < 15A). This pin can be driven directly from logic as shown in Figure 5. Diode D1 in Figure 5 reduces the start delay but allows CSS to ramp up slowly providing the soft-start function. This diode can be deleted if soft-start is not needed.
3.3V OR 5V D1 CSS CSS
1627 F05
RUN/SS
RUN/SS
(2)
Figure 5. RUN/SS Pin Interfacing
9
LTC1627
APPLICATIONS INFORMATION
Auxiliary Winding Control Using SYNC/FCB Pin The SYNC/FCB pin can be used as a secondary feedback input to provide a means of regulating a flyback winding output. When this pin drops below its ground referenced 0.8V threshold, continuous mode operation is forced. In continuous mode, the main and synchronous MOSFETs are switched continuously regardless of the load on the main output. Synchronous switching removes the normal limitation that power must be drawn from the inductor primary winding in order to extract power from auxiliary windings. With continuous synchronous operation power can be drawn from the auxiliary windings without regard to the primary output load. The secondary output voltage is set by the turns ratio of the transformer in conjunction with a pair of external resistors returned to the SYNC/FCB pin as shown in Figure 6. The secondary regulated voltage VSEC in Figure 6 is given by: Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC1627 circuits: VIN quiescent current and I2R losses. 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches RSW and external inductor RL. In continuous mode the average output current flowing through inductor L is "chopped" between the main switch and the synchronous switch. Thus, the series resistance looking into SW pin from L is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 - DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses, MOSFET switching losses and inductor core losses generally account for less than 2% total additional loss.
R4 VSEC N + 1 VOUT - VDIODE > 0.8V 1 + R3 where N is the turns ratio of the transformer, VOUT is the main output voltage sensed by VFB and VDIODE is the voltage drop across the Schottky diode.
( )(
)
R4 SYNC/FCB R3 LTC1627 SW L1 1:N
VSEC
+
1F VOUT
+
COUT
1627 F06
Figure 6. Secondary Output Loop Connection
Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as:
10
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LTC1627
APPLICATIONS INFORMATION
Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (ILOAD * ESR), where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steady-state value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The internal compensation provides adequate compensation for most applications. But if additional compensation is required, the ITH pin can be used for external compensation as shown in Figure 7. A second, more severe transient is caused by switching in loads with large (>1F) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 * CLOAD). Thus, a 10F capacitor would require a 250s rise time, limiting the charging current to about 130mA. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1627. These items are also illustrated graphically in the layout diagram of Figure 7. Check the following in your layout: 1. Are the signal and power grounds segregated? The LTC1627 signal ground consists of the resistive divider, the optional compensation network (RC and CC1), CSS and CC2. The power ground consists of the (-) plate of CIN, the (-) plate of COUT and Pin 4 of the LTC1627. The power ground traces should be kept short, direct and wide. The signal ground and power ground should converge to a common node in a starground configuration. 2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and signal ground.
CC2
OPTIONAL
RC
CC1 CSS
1 2 3 4
ITH RUN/SS
LTC1627 VFB GND VIN SW 6 5 L1
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 7. LTC1627 Layout Diagram
U
W
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SYNC/FCB VDR
8 OPTIONAL 7 CV
+ +
D1 R2 CIN VIN D2
+
+
VOUT COUT CB
R1
-
-
1627 F07
11
LTC1627
APPLICATIONS INFORMATION
3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 4. Keep the switching node SW away from sensitive smallsignal nodes. Design Example As a design example, assume the LTC1627 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.5A but most of the time it will be on standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 2.5V. With this information we can calculate L using equation (1),
L=
( )( )
V VOUT 1 - OUT VIN f IL 1
Substituting VOUT = 2.5V, VIN = 4.2V, IL = 200mA and f = 350kHz in equation (3) gives:
CITH 47pF 1 2 CSS 0.1F 3 4 ITH RUN/SS LTC1627 VFB GND VIN SW SYNC/FCB VDR 8 100 7 6 5 15H* R2 169k 1% C1 0.1F VOUT 2.5V 0.5A VIN 2.8V TO 4.5V BAT54S** D1 95 90 85 EFFICIENCY (%) 80 75 70 65 60 D2 C2 0.1F 55 50 45 1 * SUMIDA CD54-150 ** ZETEX BAT54S AVX TPSC107M006R0150 AVX TPSC226M016R0375
1627 F08a
+
R1 80.6k 1%
COUT 100F 6.3V
Figure 8. Single Lithium-Ion to 2.5V/0.5A Regulator
12
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L=
(
2.5V 1 - = 14.5H 350kHz 200mA 4.2V 2.5V
)(
)
A 15H inductor works well for this application. For good efficiency choose a 1A inductor with less than 0.25 series resistance. CIN will require an RMS current rating of at least 0.25A at temperature and COUT will require an ESR of less than 0.25. In most applications, the requirements for these capacitors are fairly similar. For the feedback resistors, choose R1 = 80.6k. R2 can then be calculated from equation (2) to be:
V R2 = OUT - 1 * R1 = 171k; use 169k 0.8
Figure 8 shows the complete circuit along with its efficiency curve.
(3)
VIN = 3.6V VIN = 4.2V
+
CIN 22F 16V
VOUT = 2.5V 10 100 OUTPUT CURRENT (mA) 1000
1627 F08b
LTC1627
TYPICAL APPLICATIONS U
5V Input to 3.3V/0.5A Regulator
CITH 47pF 1 2 CSS 0.1F 3 4 ITH SYNC/FCB VDR VIN SW 8 7 6 5 15H* R2 249k 1% R1 80.6k 1% VOUT 3.3V 0.5A
** AVX TPSC107M006R0150 ***
* SUMIDA CD54-150
AVX TPSC226M016R0375
RUN/SS LTC1627 VFB GND
VIN = 5V
+
+
COUT ** 100F 6.3V
CIN*** 22F 16V
1627 TA03
Double Lithium-Ion to 5V/0.5A Low Dropout Regulator
CITH 47pF 1 2 CSS 0.1F 3 4 ITH SYNC/FCB VDR VIN SW 8 7 6 5 33H* R2 422k 1% R1 80.6k 1% VOUT 5V 0.5A VIN 8.4V
** AVX TPSD107M010R0100 ***
* SUMIDA CD54-330
AVX TPSC226M016R0375
RUN/SS LTC1627 VFB GND
+
+
COUT ** 100F 10V
CIN*** 22F 16V
1627 TA04
13
LTC1627
TYPICAL APPLICATIONS
CITH 47pF 1 2 CSS 0.1F 3 4 ITH RUN/SS LTC1627 VFB GND VIN SW SYNC/FCB VDR 8 7 6 5 10H* R2 169k 1% C1 0.1F VOUT 2.5V 0.5A
* SUMIDA CD54-100 ** ZETEX BAT54S AVX TPSC107M006R0150 AVX TPSC226M016R0375
CITH 47pF 1 2 CSS 0.1F 3 4 ITH SYNC/FCB VDR VIN SW 8 7 6 5 15H*
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3.3V Input to 2.5V/0.5A Regulator
VIN = 3.3V
BAT54S** D1
+
+
CIN 22F 16V
R1 80.6k 1%
COUT 100F 6.3V
D2 C2 0.1F
1627 TA05
Single Lithium-Ion to 1.8V/0.3A Regulator
** AVX TPSC107M006R0150 ***
* SUMIDA CD54-150
AVX TPSC226M016R0375
RUN/SS LTC1627 VFB GND
VIN 4.2V R2 100k 1% R1 80.6k 1% VOUT 1.8V 0.3A
+
+
COUT ** 100F 6.3V
CIN*** 22F 16V
1627 TA01
LTC1627
PACKAGE DESCRIPTION
0.010 - 0.020 x 45 (0.254 - 0.508) 0.008 - 0.010 (0.203 - 0.254) 0- 8 TYP
0.016 - 0.050 0.406 - 1.270
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
U
Dimensions in inches (millimeters) unless otherwise noted.
S8 Package 8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 - 0.197* (4.801 - 5.004) 8 7 6 5
0.228 - 0.244 (5.791 - 6.197)
0.150 - 0.157** (3.810 - 3.988)
1 0.053 - 0.069 (1.346 - 1.752)
2
3
4
0.004 - 0.010 (0.101 - 0.254)
0.014 - 0.019 (0.355 - 0.483)
0.050 (1.270) TYP
SO8 0996
15
LTC1627
TYPICAL APPLICATIONS
Double Lithium-Ion to 2.5V/0.5A Regulator
CITH 47pF 1 2 CSS 0.1F 3 4 ITH SYNC/FCB VDR VIN SW 8 7 6 5 25H* R2 169k 1% R1 80.6k 1% VOUT 2.5V 0.5A VIN 8.4V * SUMIDA CD54-250
CITH 47pF 1 2 CSS 0.1F 3 4 ITH SYNC/FCB VDR VIN SW 8 7 6 5
* AVX TPSC226M016R0375 ** AVX TPSC107M006R0150 *** AVX TAJA226M006R
RELATED PARTS
PART NUMBER LTC1174/LTC1174-3.3 LTC1174-5 LTC1265 LT (R)1375/LT1376 LTC1435 LTC1436/LTC1436-PLL LTC1438/LTC1439 LTC1474/LTC1475 LTC1626 DESCRIPTION High Efficiency Step-Down and Inverting DC/DC Converters 1.2A, High Efficiency Step-Down DC/DC Converter 1.5A, 500kHz Step-Down Switching Regulators High Efficiency, Synchronous Step-Down Converter High Efficiency, Low Noise, Synchronous Step-Down Converters Dual, Low Noise, Synchronous Step-Down Converters Low Quiescent Current Step-Down DC/DC Converters Low Voltage, High Efficiency Step-Down DC/DC Converter COMMENTS Monolithic Switching Regulators, IOUT to 450mA, Burst Mode Operation Constant Off-Time, Monolithic, Burst Mode Operation High Frequency, Small Inductor, High Efficiency 16-Pin SO and SSOP 24-Pin Narrow SSOP Multiple Output Capability Monolithic, IOUT to 250mA, IQ = 10A, 8-Pin MSOP Monolithic, Constant Off-Time, IOUT to 600mA, Low Supply Voltage Range: 2.5V to 6V
16
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 q FAX: (408) 434-0507 q www.linear-tech.com
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** AVX TPSC107M006R0150 ***
AVX TPSC226M016R0375
RUN/SS LTC1627 VFB GND
+
+
COUT ** 100F 6.3V
CIN*** 22F 16V
1627 TA01
Dual Output 1.8V/300mA and 3.3V/100mA Application
R3 249k 1% R4 80.6k 1% VIN 8.5V ***22F 6.3V VSEC 3.3V 100mA D2 ZENER 1.8V VOUT 1.8V 0.3A COUT ** 100F 6.3V
+
RUN/SS LTC1627 VFB GND
D1 25H MBR0520LT1 1:1
+
CIN* 22F 16V
R2 100k 1% R1 80.6k 1%
+
COILTRONICS CTX25-1 MMSZ4678T1
A 10mA MIN LOAD CURRENT IS RECOMMENDED
1627 TA02
1627f LT/TP 0199 4K * PRINTED IN USA
(c) LINEAR TECHNOLOGY CORPORATION 1998

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