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High Common-Mode Voltage, Single-Supply Difference Amplifier AD8203
FEATURES
High common-mode voltage range -6 V to +30 V at a 5 V supply voltage Operating temperature range: -40C to +125C Supply voltage range: 3.5 V to 12 V Low-pass filter (1-pole or 2-pole)
FUNCTIONAL BLOCK DIAGRAM
NC
6
A1
3
A2
4
+VS
7
AD8203
100k G = x7 +IN 8 -IN 1 200k +IN A1 -IN 200k 10k
05013-001
G = x2 +IN A2 -IN 10k
5
OUT
EXCELLENT AC AND DC PERFORMANCE
2 mV voltage offset 1 ppm/C typ gain drift 80 dB CMRR min dc to 10 kHz
PLATFORMS
NC = NO CONNECT
2
GND
Transmission control Diesel injection control Engine management Adaptive suspension control Vehicle dynamics control
Figure 1.
CLAMP DIODE
INDUCTIVE 5V LOAD OUTPUT
+IN +VS NC OUT
GENERAL DESCRIPTION
BATTERY 14V 4-TERM SHUNT
The AD8203 is a single-supply difference amplifier for amplifying and low-pass filtering small differential voltages in the presence of a large common-mode voltage. The input CMV range extends from -6 V to +30 V at a typical supply voltage of 5 V. The AD8203 is offered in die and packaged form. The MSOP package is specified over a wide temperature range of -40C to +125C, while the die is specified over a wide temperature range of -40C to +150C, making the AD8203 well-suited for use in many automotive platforms. Automotive platforms demand precision components for better system control. The AD8203 provides excellent ac and dc performance, which keeps errors to a minimum in the user's system. Typical offset and gain drift in the MSOP package are 8 V/C and 1 ppm/C, respectively. The device also delivers a minimum CMRR of 80 dB from dc to 10 kHz. The AD8203 features an externally accessible 100 k resistor at the output of the preamp A1, which can be used for low-pass filter applications and for establishing gains other than 14.
AD8203
-IN GND A1 A2
POWER DEVICE
COMMON
NC = NO CONNECT
Figure 2. High Line Current Sensor
POWER DEVICE
5V OUTPUT
+IN
+VS
NC
OUT
BATTERY
14V 4-TERM SHUNT
AD8203
-IN GND A1 A2
CLAMP DIODE
INDUCTIVE LOAD
05013-003
COMMON
NC = NO CONNECT
Figure 3. Low Line Current Sensor
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.326.8703 (c) 2004 Analog Devices, Inc. All rights reserved.
05013-002
AD8203 TABLE OF CONTENTS
Specifications--Single Supply ......................................................... 3 Absolute Maximum Ratings............................................................ 4 ESD Caution.................................................................................. 4 Pin Configuration and Function Descriptions............................. 5 Typical Performance Characteristics ............................................. 6 Theory of Operation ........................................................................ 8 Applications....................................................................................... 9 Current Sensing ............................................................................ 9 Gain Adjustment........................................................................... 9 Gain Trim .................................................................................... 10 Low-Pass Filtering...................................................................... 10 High Line Current Sensing with LPF and Gain Adjustment................................................................. 11 Driving Charge Redistribution ADCs ..................................... 11 Outline Dimensions ....................................................................... 12 Ordering Guide .......................................................................... 12
REVISION HISTORY
10/04--Revision 0: Initial Version
Rev. 0 | Page 2 of 12
AD8203 SPECIFICATIONS--SINGLE SUPPLY
TA = operating temperature range, VS = 5 V, unless otherwise noted. Table 1.
Parameter SYSTEM GAIN Initial Error vs. Temperature VOLTAGE OFFSET Input Offset (RTI) vs. Temperature INPUT Input Impedance Differential Common-Mode CMV CMRR1 Conditions Min AD8203 MSOP Typ Max 14 0.02 VOUT 4.8 V dc -0.3 1 VCM = 0.15 V; 25C -40C to +125C -40C to +150C -2 -20 +0.3 25 +2 +20 -0.3 1 -2 -10 -15 Min AD8203 Die Typ Max 14 +0.3 30 +2 +10 +15 Unit V/V % ppm/C mV V/C V/C
+2
+0.3 +5
Continuous VCM = 0 V to 10 V f = DC f = 1 kHz f = 10 kHz2
260 130 -6 82 82 80
320 160
380 190 +30
260 130 -6 82 82 80
320 160
380 190 +30
k k V dB dB dB
PREAMPLIFIER Gain Gain Error Output Voltage Range Output Resistance OUTPUT BUFFER Gain Gain Error Output Voltage Range Input Bias Current Output Resistance DYNAMIC RESPONSE System Bandwidth Slew Rate NOISE 0.1 Hz to 10 Hz Spectral Density, 1 kHz (RTI) POWER SUPPLY Operating Range Quiescent Current vs. Temperature PSRR TEMPERATURE RANGE For Specified Performance
7 -0.3 0.02 97 +0.3 4.8 103 -0.3 0.02 97
7 +0.3 4.8 103
100 2
100 2
V/V % V k V/V % V nA kHz V/s V p-p nV/Hz
0.02 VOUT 4.8 V dc
-0.3 0.02 40 2
+0.3 4.8
-0.3 0.02 40 2 40 60 0.33 10 300
+0.3 4.8
VIN = 0.01 V p-p, VOUT = 0.14 V p-p VIN = 0.28 V, VOUT = 4 V Step
40
60 0.33 10 300
3.5 VO = 0.1 V dc VS = 3.5 V to 12 V 75 -40 0.25 83
12 1.0
3.5 0.25 75 83
12 1.0
V mA dB
+125
-40
+150
C
1 2
Source imbalance < 2 . The AD8203 preamplifier exceeds 80 dB CMRR at 10 kHz. However, since the signal is available only by way of a 100 k resistor, even the small amount of pin-to-pin capacitance between Pins 1, 8 and 3, 4 may couple an input common-mode signal larger than the greatly attenuated preamplifier output. The effect of pin-to-pin coupling may be neglected in all applications by using filter capacitors at Node 3.
Rev. 0 | Page 3 of 12
AD8203 ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Supply Voltage Transient Input Voltage (400 ms) Continuous Input Voltage (Common Mode) Reversed Supply Voltage Protection Operating Temperature Range Die MSOP Storage Temperature Output Short-Circuit Duration Lead Temperature Range (Soldering 10 sec) Rating 12.5 V 44 V 35 V 0.3 V -40C to +150C -40C to +125C -65C to +150C Indefinite 300C
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
Rev. 0 | Page 4 of 12
AD8203 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
-IN 1 GND 2 A1 3
8
+IN +VS
1036m +VS
05013-004
AD8203
7
6 NC TOP VIEW A2 4 (Not to Scale) 5 OUT
NC = NO CONNECT
Figure 4. Pin Configuration
OUT +IN
Table 3. Pin Function Descriptions
Pin No. 1 2 3 4 5 6 7 8 Mnemonic -IN GND A1 A2 OUT NC +VS +IN X -409.0 -244.6 +229.4 +410.0 +410.0 NA +121.0 -409.0 Y -205.2 -413.0 -413.0 -308.6 +272.4 NA +417.0 +205.2
1048m
-IN A2
GND
A1
Figure 5. Metallization Photograph
Rev. 0 | Page 5 of 12
04981-0-005
AD8203 TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25C, VS = 5 V, VCM = 0 V, RL = 10 k, unless otherwise noted.
90 80 -5 70 60 0
COMMON-MODE VOLTAGE (V)
-55C -40C
-10
PSRR (dB)
50 40 30 20 10 0 10 100 1k FREQUENCY (Hz) 10k
05013-006
-15
+25C
-20
+125C
-25 +150C -30 3 4 5 6 7 8 9 POWER SUPPLY (V) 10 11
05013-009
100k
12
Figure 6. Power Supply Rejection Ratio vs. Frequency
25
Figure 9. Negative Common-Mode Voltage vs. Voltage Supply
40
35
COMMON-MODE VOLTAGE (V)
20
+25C
30 -55C 25 +150C 20
OUTPUT (dB)
15
10
5
05013-007
15 +125C 10 3 -40C 4 5 6 7 8 9 POWER SUPPLY (V) 10 11
05013-010
0 100
1k
10k FREQUENCY (Hz)
100k
1M
12
Figure 7. AD8203 Bandwidth
100 95 90 85
Figure 10. Positive Common-Mode Voltage vs. Voltage Supply
5.0
4.0
OUTPUT VOLTAGE (dB)
05013-008
CMRR (dB)
80 75 70 65 60 55 50 10 100 1k FREQUENCY (Hz) 10k
3.0
2.0
1.0
05013-011
100k
0 10
100 1k LOAD RESISTANCE ()
10k
Figure 8. Common-Mode Rejection Ratio vs. Frequency
Figure 11. Output Swing vs. Load Resistance
Rev. 0 | Page 6 of 12
AD8203
0
-10
OUTPUT
INF LOAD
OUTPUT MINUS SUPPLY (mV)
-20
-30 -40
10k LOAD
4
-50
INPUT
05013-012
-60 -70 3 4 5 6 7 8 9 10 SUPPLY VOLTAGE (V) 11 12
3 CH3 100mV CH4 1.0V M 20s 2.5MS/s 400NS/PT A CH3 260mV
13
Figure 12. Swing Minus Supply vs. Supply Voltage
Figure 13. Pulse Response
Rev. 0 | Page 7 of 12
05013-013
AD8203 THEORY OF OPERATION
The AD8203 consists of a preamp and buffer arranged as shown in Figure 14. Like-named resistors have equal values. The preamp incorporates a dynamic bridge (subtractor) circuit. Identical networks (within the shaded areas), consisting of RA, RB, RC, and RG, attenuate input signals applied to Pins 1 and 8. Note that when equal amplitude signals are asserted at inputs 1 and 8, and the output of A1 is equal to the common potential (i.e., zero), the two attenuators form a balanced-bridge network. When the bridge is balanced, the differential input voltage at A1, and thus its output, is zero. Any common-mode voltage applied to both inputs keeps the bridge balanced and the A1 output at zero. Because the resistor networks are carefully matched, the common-mode signal rejection approaches this ideal state. However, if the signals applied to the inputs differ, the result is a difference at the input to A1. A1 responds by adjusting its output to drive RB, by way of RG, to adjust the voltage at its inverting input until it matches the voltage at its noninverting input. By attenuating voltages at Pins 1 and 8, the amplifier inputs are held within the power supply range, even if Pin 1 and Pin 8 input levels exceed the supply, or fall below common (ground). The input network also attenuates normal (differential) mode voltages. RC and RG form an attenuator that scales A1 feedback, forcing large output signals to balance relatively small differential inputs. The resistor ratios establish the preamp gain at 7. Because the differential input signal is attenuated and then amplified to yield an overall gain of 7, Amplifier A1 operates at a higher noise gain, multiplying deficiencies such as input offset voltage and noise with respect to Pins 1 and 8.
+IN
8
To minimize these errors while extending the common-mode range, a dedicated feedback loop is employed to reduce the range of common-mode voltage applied to A1 for a given overall range at the inputs. By offsetting the range of voltage applied to the compensator, the input common-mode range is also offset to include voltages more negative than the power supply. Amplifier A3 detects the common-mode signal applied to A1 and adjusts the voltage on the matched RCM resistors to reduce the common-mode voltage range at the A1 inputs. By adjusting the common voltage of these resistors, the common-mode input range is extended while, at the same time, the normal mode signal attenuation is reduced, leading to better performance referred to input. The output of the dynamic bridge taken from A1 is connected to Pin 3 by way of a 100 k series resistor, provided for lowpass filtering and gain adjustment. The resistors in the input networks of the preamp and the buffer feedback resistors are ratio trimmed for high accuracy. The output of the preamp drives a gain-of-2 buffer amplifier, A2, implemented with carefully matched feedback resistors RF. The 2-stage system architecture of the AD8203 enables the user to incorporate a low-pass filter prior to the output buffer. By separating the gain into two stages, a full-scale, rail-to-rail signal from the preamp can be filtered at Pin 3, and a half-scale signal, resulting from filtering, can be restored to full scale by the output buffer amp. The source resistance seen by the inverting input of A2 is approximately 100 k to minimize the effects of A2's input bias current. However, this current is quite small and errors resulting from applications that mismatch the resistance are correspondingly small.
-IN
1
RA
RA 100k A1 (TRIMMED) RCM RCM A3 RF RG
3 4
A2 RF
5
RB RG RC
RB RC
AD8203
05013-014
2
COM
Figure 14. Simplified Schematic
Rev. 0 | Page 8 of 12
AD8203 APPLICATIONS
The AD8203 difference amplifier is intended for applications where it is required to extract a small differential signal in the presence of large common-mode voltages. The input resistance is nominally 320 k and the device can tolerate common-mode voltages higher than the supply voltage and lower than ground. The open collector output stage sources current to within 20 mV of ground and to within 200 mV of VS.
VCM +VS OUT
+IN +VS NC OUT
VDIFF 2 VDIFF 2
10k
10k GAIN =
AD8203
100k
20REXT REXT + 100k GAIN 20 - GAIN
REXT = 100k
-IN
GND
A1
A2
CURRENT SENSING
High Line, High Current Sensing
Basic automotive applications making use of the large commonmode range are shown in Figure 2 and Figure 3. The capability of the device to operate as an amplifier in primary battery supply circuits is shown in Figure 2; Figure 3 illustrates the ability of the device to withstand voltages below system ground.
NC = NO CONNECT REXT
05013-016
Figure 16. Adjusting for Gains Less than 14
Low Current Sensing
The AD8203 can also be used in low current sensing applications, such as the 4 to 20 mA current loop shown in Figure 15. In such applications, the relatively large shunt resistor can degrade the common-mode rejection. Adding a resistor of equal value on the low impedance side of the input corrects for this error.
10 1%
+IN
The overall bandwidth is unaffected by changes in gain by using this method, although there may be a small offset voltage due to the imbalance in source resistances at the input to the buffer. In many cases this can be ignored, but if desired, it can be nulled by inserting a resistor equal to 100 k minus the parallel sum of REXT and 100 k, in series with Pin 4. For example, with REXT = 100 k (yielding a composite gain of x7), the optional offset nulling resistor is 50 k.
5V OUTPUT
+VS NC OUT
Gains Greater than 14
Connecting a resistor from the output of the buffer amplifier to its noninverting input, as shown in Figure 17, increases the gain. The gain is now multiplied by the factor REXT/(REXT - 100 k); for example, it is doubled for REXT = 200 k. Overall gains as high as 50 are achievable in this way. Note that the accuracy of the gain becomes critically dependent on the resistor value at high gains. Also, the effective input offset voltage at Pin 1 and Pin 8 (about six times the actual offset of A1) limits the part's use in high gain, dc-coupled applications.
+VS OUT
+
10 1%
AD8203
-IN GND A1 A2
NC = NO CONNECT
Figure 15. 4 to 20 mA Current Loop Receiver
05013-015
+IN
+VS
NC
OUT
GAIN ADJUSTMENT
The default gain of the preamplifier and buffer are x7 and x2, respectively, resulting in a composite gain of x14. With the addition of external resistor(s) or trimmer(s), the gain may be lowered, raised, or finely calibrated.
VCM
VDIFF 2 VDIFF 2
10k
10k GAIN = REXT
AD8203
100k
20REXT REXT - 100k GAIN GAIN - 20
REXT = 100k
-IN
GND
A1
A2
Since the preamplifier has an output resistance of 100 k, an external resistor connected from Pins 3 and 4 to GND decreases the gain by a factor REXT/(100 k + REXT) (see Figure 16).
NC = NO CONNECT
Figure 17. Adjusting for Gains Greater than 14
Rev. 0 | Page 9 of 12
05013-017
Gains Less than 14
AD8203
GAIN TRIM
Figure 18 shows a method for incremental gain trimming by using a trim potentiometer and external resistor REXT. The following approximation is useful for small gain ranges. G (10 M / REXT)% Thus, the adjustment range is 2% for REXT = 5 M; 10% for REXT = 1 M, and so on.
5V OUT
+IN +IN +VS NC OUT +VS NC OUT
Low-pass filters can be implemented in several ways by using the features provided by the AD8203. In the simplest case, a single-pole filter (20 dB/decade) is formed when the output of A1 is connected to the input of A2 via the internal 100 k resistor by strapping Pins 3 and 4 and a capacitor added from this node to ground, as shown in Figure 19. If a resistor is added across the capacitor to lower the gain, the corner frequency increases; it should be calculated using the parallel sum of the resistor and 100 k.
5V OUTPUT VDIFF 2
VDIFF 2
AD8203
VCM VDIFF 2
-IN GND A1 A2
AD8203
VCM VDIFF 2
-IN GND A1 A2
FC =
1 2C105
C IN FARADS
REXT
GAIN TRIM 20k MIN C
05013-018 05013-019
NC = NO CONNECT
NC = NO CONNECT
Figure 18. Incremental Gain Trim
Figure 19. Single-Pole, Low-Pass Filter Using the Internal 100 k Signal
Internal Signal Overload Considerations
When configuring gain for values other than 14, the maximum input voltage with respect to the supply voltage and ground must be considered, since either the preamplifier or the output buffer reaches its full-scale output (approximately VS - 0.2 V) with large differential input voltages. The input of the AD8203 is limited to (VS - 0.2) / 7 for overall gains 7, since the preamplifier, with its fixed gain of x7, reaches its full-scale output before the output buffer. For gains greater than 7, the swing at the buffer output reaches its full scale first and limits the AD8203 input to (VS - 0.2) / G, where G is the overall gain.
If the gain is raised using a resistor, as shown in Figure 17, the corner frequency is lowered by the same factor as the gain is raised. Thus, using a resistor of 200 k (for which the gain would be doubled), the corner frequency is now 0.796 Hz F (0.039 F for a 20 Hz corner frequency.)
5V OUT
+IN +VS NC OUT
VDIFF 2
AD8203
VCM VDIFF 2
-IN GND A1 A2
C
LOW-PASS FILTERING
In many transducer applications, it is necessary to filter the signal to remove spurious high frequency components including noise, or to extract the mean value of a fluctuating signal with a peak-to-average ratio (PAR) greater than unity. For example, a full-wave rectified sinusoid has a PAR of 1.57, a raised cosine has a PAR of 2, and a half-wave sinusoid has a PAR of 3.14. Signals having large spikes may have PARs of 10 or more. When implementing a filter, the PAR should be considered so that the output of the AD8203 preamplifier (A1) does not clip before A2, since this nonlinearity would be averaged and appear as an error at the output. To avoid this error, both amplifiers should be made to clip at the same time. This condition is achieved when the PAR is no greater than the gain of the second amplifier (2 for the default configuration). For example, if a PAR of 5 is expected, the gain of A2 should be increased to 5.
255k C NC = NO CONNECT
005013-020
FC = 1Hz - F
Figure 20. 2-Pole, Low-Pass Filter
A 2-pole filter (with a roll-off of 40 dB/decade) can be implemented using the connections shown in Figure 20. This is a Sallen-Key form based on a x2 amplifier. It is useful to remember that a 2-pole filter with a corner frequency f2 and a 1-pole filter with a corner at f1 have the same attenuation at the frequency (f22/f1). The attenuation at that frequency is 40 log (f2/f1), which is illustrated in Figure 21. Using the standard resistor value shown and equal capacitors (Figure 20), the corner frequency is conveniently scaled at 1 Hz F (0.05 F for a 20 Hz corner). A maximally flat response occurs when the resistor is lowered to 196 k and the scaling is then 1.145 Hz F. The output offset is raised by approximately 5 mV (equivalent to 250 V at the input pins).
Rev. 0 | Page 10 of 12
AD8203
FREQUENCY
ATTENUATION
40dB/DECADE 20dB/DECADE
by a 1-pole, low-pass filter, here set with a corner frequency of 3.6 Hz, which provides about 30 dB of attenuation at 100 Hz. A higher rate of attenuation can be obtained using a 2-pole filter with fC = 20 Hz, as shown in Figure 23. Although this circuit uses two separate capacitors, the total capacitance is less than half that needed for the 1-pole filter.
INDUCTIVE 5V LOAD OUTPUT
+IN +VS NC OUT
40LOG (f2/f1)
CLAMP DIODE
05013-021
A 1-POLE FILTER, CORNER f1, AND A 2-POLE FILTER, CORNER f2, HAVE THE SAME ATTENUATION -40LOG (f2/f1) AT FREQUENCY f22/f1 f1 f2 f22/f1
BATTERY
14V 4-TERM SHUNT
432k
AD8203
-IN GND A1 A2
C 50k
POWER DEVICE 127k C NC = NO CONNECT COMMON fC = 1Hz F (0.05F FOR fC = 20Hz)
05013-023
Figure 21. Comparative Responses of 1-Pole and 2-Pole Low-Pass Filters
HIGH LINE CURRENT SENSING WITH LPF AND GAIN ADJUSTMENT
Figure 22 is another refinement of Figure 2, including gain adjustment and low-pass filtering.
CLAMP DIODE INDUCTIVE 5V LOAD OUT 4V/AMP
NC OUT
Figure 23. 2-Pole Low-Pass Filter
DRIVING CHARGE REDISTRIBUTION ADCS
When driving CMOS ADCs such as those embedded in popular microcontrollers, the charge injection (Q) can cause a significant deflection in the output voltage of the AD8203. Though generally of short duration, this deflection may persist until after the sample period of the ADC has expired due to the relatively high open-loop output impedance of the AD8203. Including an R-C network in the output can significantly reduce the effect. The capacitor helps to absorb the transient charge, effectively lowering the high frequency output impedance of the AD8203. For these applications, the output signal should be taken from the midpoint of the RLAG - CLAG combination as shown in Figure 24. Since the perturbations from the analog-to-digital converter are small, the output impedance of the AD8203 appears to be low. The transient response, therefore, has a time constant governed by the product of the two LAG components, CLAG x RLAG. For the values shown in Figure 24, this time constant is programmed at approximately 10 s. Therefore, if samples are taken at several tens of microseconds or more, there are negligible charge stack-up.
5V
4 7
+IN
+VS
BATTERY
14V 4-TERM SHUNT
191k
AD8203
20k
-IN GND A1 A2
POWER DEVICE
VOS/IB NULL C
NC = NO CONNECT
COMMON
5% CALIBRATION RANGE fC = 0.796Hz F (0.22F FOR fC = 3.6Hz)
Figure 22. High Line Current Sensor Interface; Gain = x40, Single-Pole, Low-Pass Filter
A power device that is either on or off controls the current in the load. The average current is proportional to the duty cycle of the input pulse and is sensed by a small value resistor. The average differential voltage across the shunt is typically 100 mV, although its peak value is higher by an amount that depends on the inductance of the load and the control frequency. The common-mode voltage, on the other hand, extends from roughly 1 V above ground for the on condition to about 1.5 V above the battery voltage in the off condition. The conduction of the clamping diode regulates the common-mode potential applied to the device. For example, a battery spike of 20 V may result in an applied common-mode potential of 21.5 V to the input of the devices. To produce a full-scale output of 4 V, a gain x40 is used, adjustable by 5% to absorb the tolerance in the shunt. There is sufficient headroom to allow 10% overrange (to 4.4 V). The roughly triangular voltage across the sense resistor is averaged
05013-022
+IN
AD8203
A2
5
RLAG 1k CLAG 0.01F
MICROPROCESSOR A/D
-IN
10k
10k
2
05013-024
Figure 24. Recommended Circuit for Driving CMOS A/D
Rev. 0 | Page 11 of 12
AD8203 OUTLINE DIMENSIONS
3.00 BSC
8
5
3.00 BSC
4
4.90 BSC
PIN 1 0.65 BSC 1.10 MAX 8 0 0.80 0.60 0.40
0.15 0.00 0.38 0.22 COPLANARITY 0.10
0.23 0.08 SEATING PLANE
COMPLIANT TO JEDEC STANDARDS MO-187AA
Figure 25. 8-Lead Mini Small Outline Package [MSOP] (RM-8) Dimensions shown in millimeters
ORDERING GUIDE
Model AD8203YRMZ1 AD8203YRMZ-REEL1 AD8203YRMZ-REEL71 AD8203YCSURF Temperature Package -40C to +125C -40C to +125C -40C to +125C Package Description Mini Small Outline Package (MSOP) Mini Small Outline Package (MSOP) Mini Small Outline Package (MSOP) Die Package Outline RM-8 RM-8 RM-8 Branding JXY JXY JXY
1
Z = Pb-free part.
(c) 2004 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D05013-0-10/04(0)
Rev. 0 | Page 12 of 12
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