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| a FEATURES Rail-to-Rail Output Swing Single-Supply Operation: +3 V to +36 V Low Offset Voltage: 300 V Gain Bandwidth Product: 75 kHz High Open-Loop Gain: 1000 V/mV Unity-Gain Stable Low Supply Current/Per Amplifier: 150 A max APPLICATIONS Battery Operated Instrumentation Servo Amplifiers Actuator Drives Sensor Conditioners Power Supply Control OUT A -IN A +IN A V+ +IN B OUT A -IN A +IN A V- 1 2 3 4 Dual/Quad Rail-to-Rail Operational Amplifiers OP295/OP495 PIN CONNECTIONS 8-Lead Narrow-Body SO (S Suffix) 8 7 V+ OUT B -IN B +IN B +IN A V- 3 4 6 -IN B +IN B OUT A -IN A 8-Lead Epoxy DIP (P Suffix) 1 2 8 7 V+ OUT B OP295 6 5 OP295 5 14-Lead Epoxy DIP (P Suffix) 1 2 3 4 5 6 7 14 OUT D 13 -IN D 12 +IN D 16-Lead SO (300 Mil) (S Suffix) OUT A -IN A +IN A V+ +IN B -IN B OUT B NC 1 2 3 4 5 6 7 8 16 OUT D 15 -IN D 14 +IN D 13 V- OP495 11 V- OP495 12 +IN C 11 -IN C 10 +IN C 9 8 -IN C OUT C GENERAL DESCRIPTION -IN B OUT B 10 OUT C 9 NC Rail-to-rail output swing combined with dc accuracy are the key features of the OP495 quad and OP295 dual CBCMOS operational amplifiers. By using a bipolar front end, lower noise and higher accuracy than that of CMOS designs has been achieved. Both input and output ranges include the negative supply, providing the user "zero-in/zero-out" capability. For users of 3.3 volt systems such as lithium batteries, the OP295/OP495 is specified for three volt operation. Maximum offset voltage is specified at 300 V for +5 volt operation, and the open-loop gain is a minimum of 1000 V/mV. This yields performance that can be used to implement high accuracy systems, even in single supply designs. The ability to swing rail-to-rail and supply +15 mA to the load makes the OP295/OP495 an ideal driver for power transistors and "H" bridges. This allows designs to achieve higher efficiencies and to transfer more power to the load than previously possible without the use of discrete components. For applications NC = NO CONNECT that require driving inductive loads, such as transformers, increases in efficiency are also possible. Stability while driving capacitive loads is another benefit of this design over CMOS rail-to-rail amplifiers. This is useful for driving coax cable or large FET transistors. The OP295/OP495 is stable with loads in excess of 300 pF. The OP295 and OP495 are specified over the extended industrial (-40C to +125C) temperature range. OP295s are available in 8-pin plastic and ceramic DIP plus SO-8 surface mount packages. OP495s are available in 14-pin plastic and SO-16 surface mount packages. Contact your local sales office for MIL-STD-883 data sheet. REV. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. (c) Analog Devices, Inc., 1995 One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703 OP295/OP495-SPECIFICATIONS ELECTRICAL CHARACTERISTICS (@ V = +5.0 V, V S CM = +2.5 V, TA = +25 C unless otherwise noted) Min Typ 30 8 1 0 90 1000 500 110 10,000 1 5 Max 300 800 20 30 3 5 +4.0 Units V V nA nA nA nA V dB V/mV V/mV V/C V V V mV mV mV mA dB 150 0.03 75 86 1.5 51 <0.1 dB A V/s kHz Degrees V p-p nV/Hz pA/Hz Parameter INPUT CHARACTERISTICS Offset Voltage Input Bias Current Input Offset Current Input Voltage Range Common-Mode Rejection Ratio Large Signal Voltage Gain Offset Voltage Drift OUTPUT CHARACTERISTICS Output Voltage Swing High Output Voltage Swing Low Output Current POWER SUPPLY Power Supply Rejection Ratio Supply Current Per Amplifier DYNAMIC PERFORMANCE Skew Rate Gain Bandwidth Product Phase Margin NOISE PERFORMANCE Voltage Noise Voltage Noise Density Current Noise Density Symbol VOS IB IOS VCM CMRR AVO VOS/T VOH VOL IOUT PSRR ISY SR GBP O en p-p en in Conditions -40C TA +125C -40C TA +125C -40C TA +125C 0 V VCM 4.0 V, -40C TA +125C RL = 10 k, 0.005 VOUT 4.0 V RL = 10 k, -40C TA +125C RL = 100 k to GND RL = 10 k to GND IOUT = 1 mA, -40C TA +125C RL = 100 k to GND RL = 10 k to GND IOUT = 1 mA, -40C TA +125C 1.5 V VS 15 V 1.5 V VS 15 V, -40C TA +125C VOUT = 2.5 V, RL = , -40C TA +125C RL = 10 k 4.98 4.90 11 90 85 5.0 4.94 4.7 0.7 0.7 90 18 110 2 2 0.1 Hz to 10 Hz f = 1 kHz f = 1 kHz Specifications subject to change without notice. ELECTRICAL CHARACTERISTICS (@ V = +3.0 V, V S CM = +1.5 V, TA = +25 C unless otherwise noted) Min Typ 30 8 1 Max 500 20 3 +2.0 Units V nA nA V dB V/mV V/C V mV dB 150 0.03 75 85 1.6 53 <0.1 dB A V/s kHz Degrees V p-p nV/Hz pA/Hz Parameter INPUT CHARACTERISTICS Offset Voltage Input Bias Current Input Offset Current Input Voltage Range Common-Mode Rejection Ratio Large Voltage Gain Offset Voltage Drift OUTPUT CHARACTERISTICS Output Voltage Swing High Output Voltage Swing Low POWER SUPPLY Power Supply Rejection Ratio Supply Current Per Amplifier DYNAMIC PERFORMANCE Slew Rate Gain Bandwidth Product Phase Margin NOISE PERFORMANCE Voltage Noise Voltage Noise Density Current Noise Density Symbol VOS IB IOS VCM CMRR AVO VOS/T VOH VOL PSRR ISY SR GBP O en p-p en in Conditions 0 V VCM 2.0 V, -40C TA +125C RL = 10 k 0 90 110 750 1 RL = 10 k to GND RL = 10 k to GND 1.5 V VS 15 V 1.5 V VS 15 V, -40C TA +125C VOUT = 1.5 V, RL = , -40C TA +125C RL = 10 k 2.9 0.7 90 85 110 2 0.1 Hz to 10 Hz f = 1 kHz f = 1 kHz Specifications subject to change without notice. -2- REV. B OP295/OP495 ELECTRICAL CHARACTERISTICS (@ V = 15.0 V, T = +25 C unless otherwise noted) S A Parameter INPUT CHARACTERISTICS Offset Voltage Input Bias Current Input Offset Current Input Voltage Range Common-Mode Rejection Ratio Large Signal Voltage Gain Offset Voltage Drift OUTPUT CHARACTERISTICS Output Voltage Swing High Output Voltage Swing Low Output Current POWER SUPPLY Power Supply Rejection Ratio Supply Current Supply Voltage Range DYNAMIC PERFORMANCE Slew Rate Gain Bandwidth Product Phase Margin NOISE PERFORMANCE Voltage Noise Voltage Noise Density Current Noise Density Symbol VOS IB IOS VCM CMRR AVO VOS/T VOH VOL IOUT PSRR ISY VS SR GBP O en p-p en in Conditions Min Typ 30 7 1 Max 300 800 20 30 3 5 +13.5 Units V V nA nA nA nA V dB V/mV V/C V V V V mA dB dB -40C TA +125C VCM = 0 V VCM = 0 V, -40C TA +125C VCM = 0 V VCM = 0 V, -40C TA +125C -15.0 V VCM +13.5 V, -40C TA +125C RL = 10 k -15 90 1000 110 4000 1 RL = 100 k to GND RL = 10 k to GND RL = 100 k to GND RL = 10 k to GND 14.95 14.80 15 90 85 25 110 -14.95 -14.85 VS = 1.5 V to 15 V VS = 1.5 V to 15 V, -40C TA +125C VO = 0 V, RL = , VS = 18 V, -40C TA +125C +3 ( 1.5) RL = 10 k 0.03 85 83 1.25 45 <0.1 175 +36 ( 18) A V V/s kHz Degrees V p-p nV/Hz pA/Hz 0.1 Hz to 10 Hz f =1 kHz f = 1 kHz Specifications subject to change without notice. WAFER TEST LIMITS (@ V = +5.0 V, V S CM = 2.5 V, TA = +25 C unless otherwise noted) Conditions Limit 300 20 2 0 to +4 90 90 1000 4.9 150 Units V max nA max nA max V min dB min V/V V/mV min V min A max Parameter Offset Voltage Input Bias Current Input Offset Current Input Voltage Range1 Common-Mode Rejection Ratio Power Supply Rejection Ratio Large Signal Voltage Gain Output Voltage Swing High Supply Current Per Amplifier Symbol Vos IB IOS VCM CMRR PSRR AVO VOH ISY 0 V VCM 4 V 1.5 V VS 15 V RL = 10 k RL = 10 k VOUT = 2.5 V, RL = NOTES Electrical tests and wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed for standard product dice. Consult factory to negotiate specifications based on dice lot qualifications through sample lot assembly and testing. 1 Guaranteed by CMRR test. ORDERING GUIDE Model Temperature Range Package Description 8-Pin Plastic DIP 8-Pin SOIC DICE Package Option N-8 SO-8 Model Temperature Range Package Description Package Option OP295GP -40C to +125C OP295GS -40C to +125C OP295GBC +25C OP495GP -40C to +125C OP495GS -40C to +125C OP495GBC +25C 14-Pin Plastic DIP N-14 16-Pin SOL R-16 DICE REV. B -3- OP295/OP495 ABSOLUTE MAXIMUM RATINGS 1 DICE CHARACTERISTICS Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V Input Voltage2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V Differential Input Voltage2. . . . . . . . . . . . . . . . . . . . . . . +36 V Output Short-Circuit Duration . . . . . . . . . . . . . . . . . Indefinite Storage Temperature Range P, S Package . . . . . . . . . . . . . . . . . . . . . . . . -65C to +150C Operating Temperature Range OP295G, OP495G . . . . . . . . . . . . . . . . . . . -40C to +125C Junction Temperature Range P, S Package . . . . . . . . . . . . . . . . . . . . . . . . -65C to +150C Lead Temperature Range (Soldering, 60 Sec) . . . . . . . +300C Package Type 8-Pin Plastic DIP (P) 8-Pin SOIC (S) 14-Pin Plastic DIP (P) 16-Pin SO (S) JA 3 JC Unit C/W C/W C/W C/W OP295 Die Size 0.066 x 0.080 inch, 5,280 sq. mils. Substrate (Die Backside) Is Connected to V+. Transistor Count, 74. 103 158 83 98 43 43 39 30 NOTES 1 Absolute maximum ratings apply to both DICE and packaged parts, unless otherwise noted. 2 For supply voltages less than 18 V, the absolute maximum input voltage is equal to the supply voltage. 3 JA is specified for the worst case conditions, i.e., JA is specified for device in socket for cerdip, P-DIP, and LCC packages; JA is specified for device soldered in circuit board for SOIC package. OP495 Die Size 0.113 x 0.083 inch, 9,380 sq. mils. Substrate (Die Backside) Is Connected to V+. Transistor Count, 196. Typical Characteristics 140 SUPPLY CURRENT PER AMPLIFIER - A + OUTPUT SWING - Volts 15.2 15.0 14.8 14.6 14.4 14.2 VS = 15V RL = 100k RL = 10k 120 100 VS = +36V VS = +5V RL = 2k 80 VS = +3V 60 - OUTPUT SWING - Volts -14.4 -14.6 -14.8 -15.0 -15.2 -50 -25 0 25 50 TEMPERATURE - C 75 100 RL = 2k R L = 10k RL = 100k 40 20 -50 -25 0 25 50 75 100 TEMPERATURE - C Supply Current Per Amplifier vs. Temperature Output Voltage Swing vs. Temperature -4- REV. B Typical Characteristics-OP295/OP495 3.10 VS = +3V 5.10 VS = +5V OUTPUT VOLTAGE SWING - Volts 3.00 RL = 100k 2.90 RL = 10k OUTPUT VOLTAGE SWING - Volts 5.00 RL = 100k 4.90 RL = 10k 2.80 4.80 2.70 RL = 2k 2.60 4.70 RL = 2k 4.60 2.50 -50 -25 0 25 50 75 100 4.50 -50 -25 0 25 50 75 100 TEMPERATURE - C TEMPERATURE - C Output Voltage Swing vs. Temperature Output Voltage Swing vs. Temperature 200 BASED ON 600 OP AMPS 175 150 VS = +5V TA = +25C 500 BASED ON 1200 OP AMPS 450 400 350 VS = +5V TA = +25C 125 UNITS 300 100 75 50 UNITS 250 200 150 100 25 0 -250 -200 -150 -100 50 0 -100 -50 0 50 100 150 200 INPUT OFFSET VOLTAGE - V 250 300 -50 0 50 100 150 200 250 INPUT OFFSET VOLTAGE - V OP295 Input Offset (VOS) Distribution OP495 Input Offset (VOS) Distribution 250 BASED ON 600 OP AMPS 225 200 175 150 UNITS 500 BASED ON 1200 OP AMPS VS = +5V -40 T A +85C 450 400 350 300 UNITS 250 200 150 100 50 0 VS = +5V -40 T A +85C 125 100 75 50 25 0 0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 TC - V OS - V/C 0 0.4 0.8 1.2 1.6 2.0 TC - V OS - V/C 2.4 2.8 3.2 OP295 TC-VOS Distribution OP495 TC-VOS Distribution REV. B -5- OP295/OP495-Typical Characteristics 20 VS = +5V 16 100 VS = 15V VO = 10V INPUT BIAS CURRENT - nA 12 OPEN-LOOP GAIN - V/V RL = 100k 10 8 RL = 10k 4 RL = 2k 0 -50 -25 0 25 50 TEMPERATURE - C 75 100 1 -50 -25 0 25 50 75 100 TEMPERATURE - C Input Bias Current vs. Temperature Open-Loop Gain vs. Temperature 40 SOURCE 35 12 VS = +5V VO = +4V 10 OUTPUT CURRENT - mA SINK 25 20 SINK 15 10 5 0 -50 SOURCE VS = 15V OPEN-LOOP GAIN - V/V 30 8 RL = 100k 6 RL = 10k 4 RL = 2k 2 VS = +5V -25 0 25 50 75 100 0 -50 -25 0 25 50 75 100 TEMPERATURE - C TEMPERATURE - C Output Current vs. Temperature Open-Loop Gain vs. Temperature OUTPUT VOLTAGE TO RAIL 1V 100mV SOURCE VS = +5V TA = +25C SINK 10mV 1mV 100V 1A 10A 100A 1mA 10mA LOAD CURRENT Output Voltage to Supply Rail vs. Load Current -6- REV. B OP295/OP495 APPLICATIONS Rail-to-Rail Applications Information 0.1F The OP295/OP495 has a wide common-mode input range extending from ground to within about 800 mV of the positive supply. There is a tendency to use the OP295/OP495 in buffer applications where the input voltage could exceed the commonmode input range. This may initially appear to work because of the high input range and rail-to-rail output range. But above the common-mode input range the amplifier is, of course, highly nonlinear. For this reason it is always required that there be some minimal amount of gain when rail-to-rail output swing is desired. Based on the input common-mode range this gain should be at least 1.2. Low Drop-Out Reference LED R1 Q2 2N3906 3 5 10 F VIN 2 Q1 1 MAT- 03 Q2 7 6 R5 10k 1 R6 10 2 R7 510 C1 1500pF R4 R8 100 3 8 C2 10F VOUT OP295/ OP495 4 R2 27k R3 The OP295/OP495 can be used to gain up a 2.5 V or other low voltage reference to 4.5 volts for use with high resolution A/D converters that operate from +5 volt only supplies. The circuit in Figure 1 will supply up to 10 mA. Its no-load drop-out voltage is only 20 mV. This circuit will supply over 3.5 mA with a +5 volt supply. 16k +5V 0.001F +5V 20k 2 REF43 4 6 10 VOUT = 4.5V Figure 2. Low Noise Single Supply Preamplifier 1/2 OP295/ OP495 1 TO 10F Figure 1. 4.5 Volt, Low Drop-Out Reference Low Noise, Single Supply Preamplifier Most single supply op amps are designed to draw low supply current, at the expense of having higher voltage noise. This tradeoff may be necessary because the system must be powered by a battery. However, this condition is worsened because all circuit resistances tend to be higher; as a result, in addition to the op amp's voltage noise, Johnson noise (resistor thermal noise) is also a significant contributor to the total noise of the system. The choice of monolithic op amps that combine the characteristics of low noise and single supply operation is rather limited. Most single supply op amps have noise on the order of 30 nV/Hz to 60 nV/Hz and single supply amplifiers with noise below 5 nV/Hz do not exist. In order to achieve both low noise and low supply voltage operation, discrete designs may provide the best solution. The circuit on Figure 2 uses the OP295/OP495 rail-to-rail amplifier and a matched PNP transistor pair--the MAT03--to achieve zero-in/ zero-out single supply operation with an input voltage noise of 3.1 nV/Hz at 100 Hz. R5 and R6 set the gain of 1000, making this circuit ideal for maximizing dynamic range when amplifying low level signals in single supply applications. The OP295/OP495 provides rail-to-rail output swings, allowing this circuit to operate with 0 to 5 volt outputs. Only half of the OP295/OP495 is used, leaving the other uncommitted op amp for use elsewhere. The input noise is controlled by the MAT03 transistor pair and the collector current level. Increasing the collector current reduces the voltage noise. This particular circuit was tested with 1.85 mA and 0.5 mA of current. Under these two cases, the input voltage noise was 3.1 nV/Hz and 10 nV/Hz, respectively. The high collector currents do lead to a tradeoff in supply current, bias current, and current noise. All of these parameters will increase with increasing collector current. For example, typically the MAT03 has an hFE = 165. This leads to bias currents of 11 A and 3 A, respectively. Based on the high bias currents, this circuit is best suited for applications with low source impedance such as magnetic pickups or low impedance strain gages. Furthermore, a high source impedance will degrade the noise performance. For example, a 1 k resistor generates 4 nV/Hz of broadband noise, which is already greater than the noise of the preamp. The collector current is set by R1 in combination with the LED and Q2. The LED is a 1.6 V "Zener" that has a temperature coefficient close to that of Q2's base-emitter junction, which provides a constant 1.0 V drop across R1. With R1 equal to 270 , the tail current is 3.7 mA and the collector current is half that, or 1.85 mA. The value of R1 can be altered to adjust the collector current. Whenever R1 is changed, R3 and R4 should also be adjusted. To maintain a common-mode input range that includes ground, the collectors of the Q1 and Q2 should not go above 0.5 V--otherwise they could saturate. Thus, R3 and R4 have to be small enough to prevent this condition. Their values and the overall performance for two different values of R1 are summarized in Table I. Lastly, the potentiometer, R8, is needed to adjust the offset voltage to null it to zero. Similar performance can be obtained using an OP90 as the output amplifier with a savings of about 185 A of supply current. However, the output swing will not include the positive rail, and the bandwidth will reduce to approximately 250 Hz. REV. B -7- OP295/OP495 Table I. Single Supply Low Noise Preamp Performance IC = 1.85 mA R1 R3, R4 en @ 100 Hz en @ 10 Hz ISY IB Bandwidth Closed-Loop Gain Driving Heavy Loads IC = 0.5 mA 1.0 k 910 8.6 nV/Hz 10.2 nV/Hz 1.3 mA 3 A 1 kHz 1000 unless this was a low distortion application such as audio. If this is used to drive inductive loads, be sure to add diode clamps to protect the bridge from inductive kickback. Direct Access Arrangement 270 200 3.15 nV/Hz 4.2 nV/Hz 4.0 mA 11 A 1 kHz 1000 The OP295/OP495 is well suited to drive loads by using a power transistor, Darlington or FET to increase the current to the load. The ability to swing to either rail can assure that the device is turned on hard. This results in more power to the load and an increase in efficiency over using standard op amps with their limited output swing. Driving power FETs is also possible with the OP295/OP495 because of its ability to drive capacitive loads of several hundred picofarads without oscillating. Without the addition of external transistors the OP295/OP495 can drive loads in excess of 15 mA with 15 or +30 volt supplies. This drive capability is somewhat decreased at lower supply voltages. At 5 volt supplies the drive current is 11 mA. Driving motors or actuators in two directions in a single supply application is often accomplished using an "H" bridge. The principle is demonstrated in Figure 3a. From a single +5 volt supply this driver is capable of driving loads from 0.8 V to 4.2 V in both directions. Figure 3b shows the voltages at the inverting and noninverting outputs of the driver. There is a small crossover glitch that is frequency dependent and would not cause problems +5V 2N2222 10k 0 VIN 2.5V 5k 1.67V 10k 10k 2N2907 2N2907 OUTPUTS OP295/OP495 can be used in a single supply Direct Access Arrangement (DAA) as is shown an in Figure 4. This figure shows a portion of a typical DM capable of operating from a single +5 volt supply and it may also work on +3 volt supplies with minor modifications. Amplifiers A2 and A3 are configured so that the transmit signal TXA is inverted by A2 and is not inverted by A3. This arrangement drives the transformer differentially so that the drive to the transformer is effectively doubled over a single amplifier arrangement. This application takes advantage of the OP295/OP495's ability to drive capacitive loads, and to save power in single supply applications. 390pF 37.4k 20k 0.1F RXA 0.0047F 3.3k 20k 475 A2 22.1k 0.1F TXA 20k 20k 0.033F 20k 750pF 1:1 A1 OP295/ OP495 OP295/ OP495 2.5V REF OP295/ OP495 A3 2N2222 Figure 4. Direct Access Arrangement A Single Supply Instrumentation Amplifier The OP295/OP495 can be configured as a single supply instrumentation amplifier as in Figure 5. For our example, VREF is set V+ equal to and VO is measured with respect to VREF. The in2 put common-mode voltage range includes ground and the output swings to both rails. V+ Figure 3a. "H" Bridge VIN 3 1/2 OP295/ OP495 1 5 8 1/2 OP295/ OP495 7 VO 6 4 2 100 90 R1 100k VREF R2 20k RG VO = 5 + R3 20k R4 100k ( 200k RG )V IN + VREF 10 0% Figure 5. Single Supply Instrumentation Amplifier 2V 2V 1ms Figure 3b. "H" Bridge Outputs Resistor RG sets the gain of the instrumentation amplifier. Minimum gain is 6 (with no RG). All resistors should be matched in absolute value as well as temperature coefficient to maximize -8- REV. B OP295/OP495 common-mode rejection performance and minimize drift. This instrumentation amplifier can operate from a supply voltage as low as 3 volts. A Single Supply RTD Thermometer Amplifier This RTD amplifier takes advantage of the rail-to-rail swing of the OP295/OP495 to achieve a high bridge voltage in spite of a low 5 V supply. The OP295/OP495 amplifier servos a constant 200 A current to the bridge. The return current drops across the parallel resistors 6.19 k and the 2.55 M, developing a voltage that is servoed to 1.235 V, which is established by the AD589 bandgap reference. The 3-wire RTD provides an equal line resistance drop in both 100 legs of the bridge, thus improving the accuracy. The AMP04 amplifies the differential bridge signal and converts it to a single-ended output. The gain is set by the series resistance of the 332 resistor plus the 50 potentiometer. The gain scales the output to produce a 4.5 V full scale. The 0.22 F capacitor to the output provides a 7 Hz low-pass filter to keep noise at a minimum. 200 10-TURNS 26.7k 0.5% ZERO ADJ +5V 26.7k 0.5% 7 3 1 8 0.22F AMP04 2 100 RTD 100 0.5% 2 1 To calibrate, immerse the thermocouple measuring junction in a 0C ice bath, adjust the 500 Zero Adjust pot to zero volts out. Then immerse the thermocouple in a 250C temperature bath or oven and adjust the Scale Adjust pot for an output voltage of 2.50 V, which is equivalent to 250C. Within this temperature range, the K-type thermocouple is quite accurate and produces a fairly linear transfer characteristic. Accuracy of 3C is achievable without linearization. Even if the battery voltage is allowed to decay to as low as 7 volts, the rail-to-rail swing allows temperature measurements to 700C. However, linearization may be necessary for temperatures above 250C where the thermocouple becomes rather nonlinear. The circuit draws just under 500 A supply current from a 9 V battery. A 5 V Only, 12-Bit DAC That Swings 0 V to 4.095 V 50 332 Figure 8 shows a complete voltage output DAC with wide output voltage swing operating off a single +5 V supply. The serial input 12-bit D/A converter is configured as a voltage output device with the 1.235 V reference feeding the current output pin (IOUT) of the DAC. The VREF which is normally the input now becomes the output. The output voltage from the DAC is the binary weighted voltage of the reference, which is gained up by the output amplifier such that the DAC has a 1 mV per bit transfer function. 6 VO 4.5V = 450C 0V = 0C 1/2 OP295/ OP495 3 5 4 +5V +5V 8 VDD RFB VREF 2 1 3 +5V R1 17.8k +5V +1.23V 3 IOUT 1.235 37.4k 8 2.55M 1% 6.19k 1% DAC8043 VO = 1 AD589 GND CLK SRI LD AD589 4 7 6 5 OP295/ OP495 2 4 R4 R2 41.2k R3 5k D (4.096V) 4096 Figure 6. Low Power RTD Amplifier A Cold Junction Compensated, Battery Powered Thermocouple Amplifier DIGITAL CONTROL TOTAL POWER DISSIPATION = 1.6mW 100k The OP295/OP495's 150 A quiescent current per amplifier consumption makes it useful for battery powered temperature measuring instruments. The K-type thermocouple terminates into an isothermal block where the terminated junctions' ambient temperatures can be continuously monitored and corrected by summing an equal but opposite thermal EMF to the amplifier, thereby canceling the error introduced by the cold junctions. 1.235V AD589 ISOTHERMAL BLOCK 1N914 ALUMEL AL 1.5M 1% 24.9k 9V 24.3k 1% 4.99k 1% 2 8 Figure 8. A 5 Volt 12-Bit DAC with 0 V to +4.095 Output Swing 4-20 mA Current Loop Transmitter Figure 9 shows a self powered 4-20 mA current loop transmitter. The entire circuit floats up from the single supply (12 V to 36 V) return. The supply current carries the signal within the 4 to 20 mA range. Thus the 4 mA establishes the baseline NULL ADJ SPAN ADJ 100k 10-TURN 6 REF02 GND 5V 3 8 220 1 2 4 4 100 +12V TO +36V 2 7.15k 1% 24.9k 1% SCALE ADJUST 1.33M 20k 10k 182k 1.21M 10-TURN 1% 1% 1 COLD JUNCTIONS 500 10-TURN 3 OP295/ OP495 4 CR CHROMEL K-TYPE THERMOCOUPLE 40.7V/C 475 1% 2.1k 1% ZERO ADJUST VO 0V = 0C 5V = 500C VIN 0 + 3V 1/2 OP295/ OP495 2N1711 4-20mA RL 100 100 1% 220pF 100k HP 5082-2800 1% Figure 7. Battery Powered, Cold-Junction Compensated Thermocouple Amplifier Figure 9. 4-20 mA Current Loop Transmitter REV. B -9- OP295/OP495 current budget with which the circuit must operate. This circuit consumes only 1.4 mA maximum quiescent current, making 2.6 mA of current available to power additional signal conditioning circuitry or to power a bridge circuit. A 3 Volt Low-Dropout Linear Voltage Regulator current limit loop. At this point A2's lower output resistance dominates the drive to the power MOSFET transistor, thereby effectively removing the A1 voltage regulation loop from the circuit. If the output current greater than 1 amp persists, the current limit loop forces a reduction of current to the load, which causes a corresponding drop in output voltage. As the output voltage drops, the current limit threshold also drops fractionally, resulting in a decreasing output current as the output voltage decreases, to the limit of less than 0.2 A at 1 V output. This "fold-back" effect reduces the power dissipation considerably during a short circuit condition, thus making the power supply far more forgiving in terms of the thermal design requirements. Small heat sinking on the power MOSFET can be tolerated. The OP295's rail-to-rail swing exacts higher gate drive to the power MOSFET, providing a fuller enhancement to the transistor. The regulator exhibits 0.2 V dropout at 500 mA of load current. At 1 amp output, the dropout voltage is typically 5.6 volts. IRF9531 S D 6V G 8 7 1N4148 A2 6 45.3k 1% 45.3k 1% 5 RSENSE 0.1 1/4W 210k 1% IO (NORM) = 0.5A IO (MAX) = 1A +5V 205k 1% VO Figure 10 shows a simple 3 V voltage regulator design. The regulator can deliver 50 mA load current while allowing a 0.2 V dropout voltage. The OP295/OP495's rail-to-rail output swing handily drives the MJE350 pass transistor without requiring special drive circuitry. At no load, its output can swing less than the pass transistor's base-emitter voltage, turning the device nearly off. At full load, and at low emitter-collector voltages, the transistor beta tends to decrease. The additional base current is easily handled by the OP295/OP495 output. The amplifier servos the output to a constant voltage, which feeds a portion of the signal to the error amplifier. Higher output current, to 100 mA, is achievable at a higher dropout voltage of 3.8 V. MJE 350 VIN 5V TO 3.2V 1 IL < 50mA VO 44.2k 1% 8 100F 1/2 OP295/ OP495 4 3 30.9k 1% 2 1000pF 1.235V 43k AD589 100k 5% 1/2 OP295/ OP495 0.01F 3 1 A1 2 Figure 10. 3 V Low Dropout Voltage Regulator Figure 11 shows the regulator's recovery characteristic when its output underwent a 20 mA to 50 mA step current change. 2V 100 1/2 4 OP295/ OP495 2 REF43 4 6 124k 1% 124k 1% 2.500V STEP CURRENT CONTROL WAVEFORM 50mA 90 Figure 12. Low Dropout, 500 mA Voltage Regulator with Fold-Back Current Limiting 20mA OUTPUT 10 0% 20mV 1ms Square Wave Oscillator The circuit in Figure 13 is a square wave oscillator (note the positive feedback). The rail-to-rail swing of the OP295/OP495 helps maintain a constant oscillation frequency even if the supply voltage varies considerably. Consider a battery powered system where the voltages are not regulated and drop over time. The rail-to-rail swing ensures that the noninverting input sees the full V+/2, rather than only a fraction of it. The constant frequency comes from the fact that the 58.7 k feedback sets up Schmitt Trigger threshold levels that are directly proportional to the supply voltage, as are the RC charge voltage levels. As a result, the RC charge time, and therefore the frequency, remains constant independent of supply voltage. The slew rate of the amplifier limits the oscillation frequency to a maximum of about 800 Hz at a +5 V supply. Single Supply Differential Speaker Driver Figure 11. Output Step Load Current Recovery Low-Dropout, 500 mA Voltage Regulator with Fold-Back Current Limiting Adding a second amplifier in the regulation loop as shown in Figure 12 provides an output current monitor as well as foldback current limiting protection. Amplifier A1 provides error amplification for the normal voltage regulation loop. As long as the output current is less than 1 ampere, amplifier A2's output swings to ground, reverse biasing the diode and effectively taking itself out of the circuit. However, as the output current exceeds 1 amp, the voltage that develops across the 0.1 sense resistor forces the amplifier A2's output to go high, forward-biasing the diode, which in turn closes the Connected as a differential speaker driver, the OP295/OP495 can deliver a minimum of 10 mA to the load. With a 600 load, the OP295/OP495 can swing close to 5 volts peak-to-peak across the load. -10- REV. B OP295/OP495 V+ 100k 58.7k 3 8 1 FREQ OUT fOSC = 1 < 350Hz @ V+ = +5V RC 1/2 OP295/ OP495 2 100k R 4 C Figure 13. Square Wave Oscillator Has Stable Frequency Regardless of Supply Changes 90.9k 10k 2.2F VIN 10k V+ 1/4 OP295/ OP495 100k SPEAKER 1/4 OP295/ OP495 20k V+ 20k 1/4 OP295/ OP495 Figure 14. Single Supply Differential Speaker Driver High Accuracy, Single-Supply, Low Power Comparator The OP295/OP495 makes an accurate open-loop comparator. With a single +5 V supply, the offset error is less than 300 V. Figure 15 shows the OP295/OP495's response time when operating open-loop with 4 mV overdrive. It exhibits a 4 ms response time at the rising edge and a 1.5 ms response time at the falling edge. 1V 100 90 INPUT (5mV OVERDRIVE @ OP295 INPUT) OUTPUT 10 0% 2V 5ms Figure 15. Open-Loop Comparator Response Time with 5 mV Overdrive OP295/OP495 SPICE MODEL Macro-Model * Node Assignments * Noninverting Input * Inverting Input * Positive Supply * Negative Supply * Output * * .SUBCKT OP295 1 2 99 50 20 * * INPUT STAGE * I1 99 4 2E-6 R1 1 6 5E3 R2 2 5 5E3 CIN 1 2 2E-12 IOS 1 2 0.5E-9 D1 5 3 DZ D2 6 3 DZ EOS 7 6 POLY (1) (31,39) 30E-6 0.024 Q1 8 54 QP Q2 9 7 4 QP R3 8 50 25.8E3 R4 9 50 25.8E3 * * GAIN STAGE * R7 10 98 270E6 G1 98 10 POLY (1) (9,8) -4.26712E-9 27.8E-6 EREF 98 0 (39, 0) 1 R5 99 39 100E3 R6 39 50 100E3 * * COMMON MODE STAGE * ECM 30 98 POLY(2) (1,39) (2,39) 0 0.5 0.5 R12 30 31 1E6 R13 31 98 100 * * OUTPUT STAGE * I2 18 50 1.59E-6 V2 99 12 DC 2.2763 Q4 10 14 50 QNA 1.0 R11 14 50 33 M3 15 10 13 13 MN L=9E-6 W=102E-6 AD=15E-10 AD=15E-10 M4 13 10 50 50 MN L=9E-6 W=50E-6 AD=75E-11 AS=75E-11 D8 10 22 DX V3 22 50 DC 6 M2 20 10 14 14 MN L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9 Q5 17 17 99 QPA 1.0 Q6 18 17 99 QPA 4.0 R8 18 99 2.2E6 Q7 18 19 99 QPA 1.0 R9 99 19 8 C2 18 99 20E-12 M6 15 12 17 99 MP L=9E-6 W=27E-6 AD=405E-12 AS=405E-12 M1 20 18 19 99 MP L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9 D4 21 18 DX V4 99 21 DC 6 R10 10 11 6E3 C3 11 20 50E-12 .MODEL QNA NPN (IS=1.19E-16 BF=253 NF=0.99 VAF=193 IKF=2.76E-3 + ISE=2.57E-13 NE=5 BR=0.4 NR=0.988 VAR=15 IKR=1.465E-4 + ISC=6.9E-16 NC=0.99 RB=2.0E3 IRB=7.73E-6 RBM=132.8 RE=4 RC=209 + CJE=2.1E-13 VJE=0.573 MJE=0.364 FC=0.5 CJC=1.64E-13 VJC=0.534 MJC=0.5 + CJS=1.37E-12 VJS=0.59 MJS=0.5 TF=0.43E-9 PTF=30) .MODEL QPA PNP (IS=5.21E-17 BF=131 NF=0.99 VAF=62 IKF=8.35E-4 + ISE=1.09E-14 NE=2.61 BR=0.5 NR=0.984 VAR=15 IKR=3.96E-5 + ISC=7.58E-16 NC=0.985 RB=1.52E3 IRB=1.67E-5 RBM=368.5 RE=6.31 RC=354.4 + CJE=1.1E-13 VJE=0.745 MJE=0.33 FC=0.5 CJC=2.37E-13 VJC=0.762 MJC=0.4 + CJS =7.11E-13 VJS=0.45 MJS=0.412 TF=1.0E-9 PTF=30) .MODEL MN NMOS (LEVEL=3 VTO=1.3 RS=0.3 RD=0.3 + TOX=8.5E-8 LD=1.48E-6 NSUB=1.53E16 UO=650 DELTA=10 VMAX=2E5 + XJ=1.75E-6 KAPPA=0.8 ETA=0.066 THETA=0.01 TPG=1 CJ=2.9E-4 PB=0.837 + MJ=0.407 CJSW=0.5E-9 MJSW=0.33) .MODEL MP PMOS (LEVEL=3 VTO=-1.1 RS=0.7 RD=0.7 + TOX=9.5E-8 LD=1.4E-6 NSUB=2.4E15 UO=650 DELTA=5.6 VMAX=1E5 + XJ=1.75E-6 KAPPA=1.7 ETA=0.71 THETA=5.9E-3 TPG=-1 CJ=1.55E-4 PB=0.56 + MJ=0.442 CJSW=0.4E-9 MJSW=0.33) .MODEL DX D(IS=1E-15) .MODEL DZ D (IS=1E-15, BV=7) .MODEL QP PNP (BF=125) .ENDS REV. B -11- OP295/OP495 OUTLINE DIMENSIONS Dimensions shown in inches and (mm) 8 Lead Plastic DIP (P Suffix) 8 8-Lead Narrow-Body SO (S Suffix) 5 8 5 0.1574 (4.00) 0.1497 (3.80) 1 4 0.070 (1.77) 0.045 (1.15) PIN 1 1 0.325 (8.25) 0.300 (7.62) 0.195 (4.95) 0.115 (2.93) 4 0.430 (10.92) 0.348 (8.84) 0.210 (5.33) MAX 0.160 (4.06) 0.115 (2.93) 0.2440 (6.20) 0.2284 (5.80) 0.015 (0.381) TYP 0.1968 (5.00) 0.1890 (4.80) 0.0098 (0.25) 0.0688 (1.75) 0.0532 (1.35) 0.0500 (1.27) BSC 0.0192 (0.49) 0.0138 (0.35) 8 0 0.0196 (0.50) x 45 0.0099 (0.25) 0.130 (3.30) MIN 0.100 (2.54) BSC SEATING PLANE 0- 15 0.015 (0.381) 0.008 (0.204) 0.0040 (0.10) 0.0098 (0.25) 0.0075 (0.19) 0.0500 (1.27) 0.0160 (0.41) 0.022 (0.558) 0.014 (0.356) 14-Lead Plastic DIP (P Suffix) 14 PIN 1 1 0.795 (20.19) 0.725 (18.42) 0.210 (5.33) MAX 0.160 (4.06) 0.115 (2.92) 0.022 (0.558) 0.014 (0.36) 0.015 (0.381) MIN 7 0.325 (8.25) 0.300 (7.62) PIN 1 1 16-Lead Wide-Body SO (S Suffix) 8 0.280 (7.11) 0.240 (6.10) 16 9 0.2992 (7.60) 0.2914 (7.40) 0.4193 (10.65) 0.3937 (10.00) 8 0.130 (3.30) MIN SEATING PLANE 0.015 (0.38) 0.008 (0.20) 15 0 0.4133 (10.50) 0.3977 (10.10) 0.1043 (2.65) 0.0926 (2.35) 0.100 (2.54) BSC 0.070 (1.77) 0.045 (1.15) 0.0291 (0.74) x 45 0.0098 (0.25) 0.0118 (0.30) 0.0040 (0.10) 0.0500 (1.27) BSC 0.0192 (0.49) 0.0138 (0.35) 0.0125 (0.32) 0.0091 (0.23) 8 0 0.0500 (1.27) 0.0157 (0.40) -12- REV. B PRINTED IN U.S.A. C1806a-10-7/95 0.280 (7.11) 0.240 (6.10) |
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